Method and receiver for receiving a binary offset carrier composite signal

ABSTRACT

A data processor selects a set of BOC correlations in accordance with a BOC correlation function for the sampling period if the primary amplitude exceeds or equals the secondary amplitude for the sampling period. The data processor selects a set of QBOC correlations in accordance with a QBOC correlation function for the sampling period if the secondary amplitude exceeds the primary amplitude for the sampling period. The data processor uses either the BOC correlation function or the QBOC correlation function, whichever with greater amplitude, at each sampling period to provide an aggregate correlation function that supports unambiguous code acquisition of the received signal.

RELATED APPLICATIONS

This is a continuation of U.S. application Ser. No. 14/310,610, filedJun. 20, 2014, which claims priority and the benefit of the filing datebased on U.S. provisional application No. 61/925,752, filed Jan. 10,2014 under 35 U.S.C. §119 (e), where the provisional application ishereby incorporated by reference herein.

FIELD OF THE INVENTION

This invention relates to a method and receiver for receiving acomposite signal.

BACKGROUND

A transmitter of a navigation satellite might transmit a compositesignal such as a multiplexed binary offset carrier signal or a binaryoffset carrier (BOC) signal. In certain prior art, the receiver may lockon a false zero crossing point of a discriminator function that resultsin a synchronization error, reduced reliability in decoding, orinstability of the demodulator in demodulating the composite signal.Thus, there is a need for a method and receiver with an unambiguouscorrelation function for reducing or minimizing phase synchronizationerror to the received signal.

SUMMARY OF THE INVENTION

In one embodiment, a method or receiver for receiving a receivedcomposite signal comprises receiving a composite signal, which comprisesa binary offset carrier (BOC) modulated signal, to extract a BOCcomponent by combining the received signal with a local BOC replica, andto derive a quadrature BOC (QBOC) component by combining the receivedsignal with a local QBOC replica. The BOC component comprises anin-phase BOC component and a quadrature-phase BOC component; the QBOCcomponent comprises in-phase QBOC component and a quadrature-phase QBOCcomponent. A first detector detects a primary amplitude of the BOCcomponent. A secondary amplitude of the QBOC component is detected. Anelectronic data processor determines whether the primary amplitudeexceeds the secondary amplitude for a sampling period. The dataprocessor selects a set of BOC correlations in accordance with a BOCcorrelation function for the sampling period if the primary amplitudeexceeds (e.g., or equals) the secondary amplitude for the samplingperiod. The data processor selects a set of QBOC correlations inaccordance with a QBOC correlation function for the sampling period ifthe secondary amplitude exceeds the primary amplitude for the samplingperiod. In one embodiment, the data processor uses either the BOCcorrelation function or the QBOC correlation function at each samplingperiod to provide an aggregate correlation function (e.g.,intermittently alternating correlation functions over multiple samplingperiods without simultaneous joint selection of the BOC and QBOCcorrelations during any single sampling period or epoch) that supportsunambiguous code acquisition of the received signal. The data processorprocesses the selected correlations (e.g., BOC or QBOC correlations) totrack code and/or carrier (e.g., carrier phase, carrier frequency, orboth) of the received composite signal for estimation of a range betweena receiver antenna and a satellite transmitter that transmits thereceived composite signal.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of one embodiment of a receiver for receivinga received composite signal that comprises a binary offset carrier (BOC)modulated signal; FIG. 1 includes FIG. 1A and FIG. 1B, collectively.

FIG. 2 is a block diagram of the digital section of the receiver of FIG.1 in greater detail than FIG. 1; FIG. 2 includes FIG. 2A and FIG. 2B,collectively.

FIG. 3 is a block diagram of the detectors (e.g., envelope detectors),decision unit, and code (CD) and carrier (CR) tracking module of FIG. 1in greater detail than FIG. 1; FIG. 3 includes FIG. 3A and FIG. 3B,collectively.

FIG. 4 is a block diagram of the detectors of FIG. 3 in greater detailthan FIG. 3.

FIG. 5 is a block diagram of the CD error estimator of FIG. 3 in greaterdetail than FIG. 3; FIG. 5 includes FIG. 5A and FIG. 5B, collectively.

FIG. 6 is a block diagram of the CR frequency error and CR phase errorestimator of FIG. 3 in greater detail than FIG. 3.

FIG. 7A is a flow chart of one embodiment of a method for demodulating areceived composite signal that comprises a binary offset carrier (BOC)modulated signal.

FIG. 7B (collectively FIG. 7B-1 and FIG. 7B-2) is a flow chart ofanother embodiment of a method for demodulating a received compositesignal that comprises a binary offset carrier (BOC) modulated signal.

FIG. 7C (collectively FIG. 7C-1 and FIG. 7C-2) is a flow chart of yetanother embodiment of a method for demodulating a received compositesignal that comprises a binary offset carrier (BOC) modulated signal.

FIG. 7D (collectively FIG. 7D-1 and FIG. 7D-2) is a flow chart of stillanother embodiment of a method for demodulating a received compositesignal that comprises a binary offset carrier (BOC) modulated signal.

FIG. 8 is a graph of a BOC correlation function (e.g., for a BOC(1,1)signal) and QBOC correlation function (e.g., for a QBOC(1,1) signal)using a closed-formula representation.

FIG. 9 is a graph of decision-directed selection (DDSel) amplitudederived from either the BOC or the QBOC signal components, and isrepresentative of an aggregate correlation function resulting fromeither BOC or QBOC correlation at each sampling period.

FIG. 10 is a flow chart of another embodiment of a method for receivinga received composite signal that comprises a binary offset carrier (BOC)modulated signal.

FIG. 11 is one embodiment of a method for receiving a composite signal.

FIG. 12 (collectively FIG. 12-1 and FIG. 12-2) is another embodiment ofa method for receiving a composite signal.

FIG. 13 (collectively FIG. 13-1 and FIG. 13-2) is yet another embodimentof a method for receiving a composite signal.

FIG. 14 is still another embodiment of a method for receiving acomposite signal.

DESCRIPTION OF THE PREFERRED EMBODIMENT

In accordance with one embodiment, FIG. 1 (collectively FIG. 1A and FIG.1B) discloses a system or receiver 11 (e.g., satellite navigationreceiver) capable of reliable and expedient code (CD) and carrier (CR)pull-in for received composite signals, such as a binary offset carrier(BOC) signal. As used in this document, “CD” shall refer to code and“CR” shall refer to the carrier of the received signal or a digitalrepresentation of one or more samples of the received signal. The codecomprises a modulating code (e.g., pseudo-random noise code modulatedwith information) that modulates the carrier. “I” shall refer to anin-phase signal, whereas “Q ” shall refer to a quadrature phase signal.The receiver 11 receives a composite signal that comprises one or moreBOC modulated signals. The received composite signal is transmitted fromone or more satellites, such as a navigation satellite, or such as aGalileo-compatible navigation satellite or Global Positioning System(GPS) satellite.

In any of the above referenced drawings of this document, any arrow orline that connects any blocks, components, modules, multiplexers,memory, data storage, accumulators, data processors, electroniccomponents, oscillators, signal generators, or other electronic orsoftware modules may comprise one or more of the following items: aphysical path of electrical signals, a physical path of anelectromagnetic signal, a logical path for data, one or more data buses,a circuit board trace, a transmission line; a link, call, communication,or data message between software modules, programs, data, or components;or transmission or reception of data messages, software instructions,modules, subroutines or components. In one embodiment, the system,method and receiver 11 disclosed in this document may comprise acomputer-implemented system, method or receiver 11 in which one or moredata processors process, store, retrieve, and otherwise manipulate datavia data buses and one or more data storage devices (e.g., accumulatorsor memory) as described in this document and the accompanying drawings.

As used in this document, “configured to, adapted to, or arranged to”mean that the data processor or receiver 11 is programmed with suitablesoftware instructions, software modules, executable code, datalibraries, and/or requisite data to execute any referenced functions,mathematical operations, logical operations, calculations,determinations, processes, methods, algorithms, subroutines, or programsthat are associated with one or more blocks set forth in FIG. 1 and/orany other drawing in this disclosure. Alternately, separately from orcumulatively with the above definition, “configured to, adapted to, orarranged to” can mean that the receiver 11 comprises one or morecomponents described herein as software modules, equivalent electronichardware modules, or both to execute any referenced functions,mathematical operations, calculations, determinations, processes,methods, algorithms, subroutine.

As used in this document, the BOC signal may include one or more of thefollowing types of signals: a BOC signal represented by a binary sinefunction, a quadrature-phase BOC (QBOC) represented by a binary cosinefunction, a time-division multiplexed BOC (TMBOC) signal (e.g., an L1Csignal of a GPS system), an alternative BOC (AltBOC) signal (e.g.; E5Aand E5B signal on Galileo system), or a composite BOC (cBOC, e.g., anE1B or E1C signal on Galileo system) signal. A simple BOC signal isdenoted as BOC (m, n) where m is f_(m)/f_(c) and n is f_(n)/f_(c), f_(m)is a first subcarrier frequency, f_(n) is the actual chip frequency, andf_(c) is the reference chipping rate of 1.023 MHz or another suitablechipping rate.

In one embodiment, the receiver 11 comprises an analog receiver portion191 coupled to a digital receiver portion 192. The analog receiverportion 191 comprises an antenna 20, an amplifier 21 and a receiverfront end 190. The digital receiver portion 192 includes that portion ofthe receiver that processes data after the analog-to-digital conversionby the analog-to-digital converter (ADC or A/D) 24. For example, thedigital receiver portion 192 can comprise an electronic data processor,a data storage device (e.g., electronic memory) and a data bus forcommunication between the electronic data processor and the data storagedevice, where software instructions and data are stored in the datastorage device and executed by the data processor to implement any ofthe blocks, components or modules (e.g., electronic modules, softwaremodules, or both) illustrated in FIG. 1. The receiver 11 may comprise alocation-determining receiver for: (a) determining a location of areceiver antenna 20, (b) a range-determining receiver for determining arange or distance between the receiver antenna 20 and a satellite (e.g.,satellite antenna) or (c) determining ranges between the receiverantenna 20 and one or more satellites.

In one embodiment, a receiver front end 190 receives an amplifiedcomposite signal (e.g., from the amplifier 21). In turn, the receiverfront end 190 is coupled to an analog-to-digital converter 24. Thereceiver front end 190 comprises a downconversion mixer 23 and a localoscillator 22. For example, the amplifier 21 comprises radio frequency(RF) or microwave amplifier (e.g., low noise amplifier) that is coupledto the antenna 20 for receiving the composite signal or received signalthat is transmitted from one or more satellites. The amplifier 21provides an amplified signal to the downconversion mixer 23 as a firstinput. The local oscillator 22 provides a signal to the downconversionmixer 23 as a second input. The downconversion mixer moves the signalspectrum of the received signal from RF to an intermediate frequency(IF) or baseband frequency. The downconversion system may includemultiple mixing, amplifying, and filtering stages, although only onestage is shown in FIG. 1A.

The output of the downconversion mixer 23 or the output of the receiverfront end 190 is coupled to an analog-to-digital converter (ADC) 24. TheADC 24 converts the analog intermediate frequency signal or analogbaseband signal to a digital signal. The digital signal comprises one ormore digital samples that are available at a sampling rate. Each samplehas a finite quantization level and each sample is capable of beingprocessed by an electronic data processing system or digital receiverportion 192.

The digital signal outputted by the ADC 24 is fed into a carrierwipe-off module 26. In one embodiment, the carrier wipe-off module 26converts the digital samples of digital signal 101 to an exact basebanddigital signal 102 representation by removing the residual CR frequency.The carrier NCO module 34 provides a local estimation of CR phase toeach digital sample 101, which is used to remove the residual CRfrequency and phase in the sample 101.

In one configuration, the a digital signal 101 inputted into the carrierwipe-off module 26 comprises a digital signal with a residual frequencycomponent (e.g., residual carrier radio frequency component) such thatthe carrier wipe-off module 26 produces exact digital baseband signals102 for input into a correlator module 130. An output of the carrierwipe-off module 26 is fed into a bank or set of correlators orcorrelator modules 130. In one embodiment, there is at least onecorrelator or correlator module 130 per received channel or carrier ofthe composite received signal, where each satellite within a set ofsatellites may transmit at least one channel or carrier. An example ofcorrelator module 130 may comprise a code wipe-off module 27 and anintegration-and-dump module 28.

In one embodiment, in the receiver 11 a correlator module 130 comprisesa CD correlator; where their multiple outputs are used to synchronizethe local CD phase, CR frequency, and CR phase estimation with thereceived samples. For example, each correlator module 130 comprises oneor more of the following modules: a CD wipe-off module 27, anintegration-and-dump module 28, a multiplier or mixer 42, and one ormore multiplexers (29, 30). As used in this document, a module maycomprise hardware, software, or both. In one embodiment, each correlatormodule 130 maximizes a correlation between the received signal with alocally generated code by synchronizing the locally generated CD phasewith the CD phase in digital sample or digital signal (101 or 102).Further, multiple locally generated CD signals (e.g., early, prompt andlate CD signals) are used to form a corresponding CD misalignment signalusing various discriminator functions.

A first signal generator 32 generates a locally generated replica of apseudo random noise code, a pseudo noise (PN) code sequence, or thelike. The first signal generator 32 has multiple outputs that are offsetin time or phase with respect to each other. As illustrated, the firstsignal generator 32 has an early output, a prompt output and a lateoutput, which are inputted to a first multiplexer 30 associated with thecode correlator module 130. “E,” “P,” and “L” shall mean early, promptand late, respectively. The early output provides an early PN code thatis advanced against the current estimated code phase by a known timeperiod (e.g., one chip); the prompt output provides a prompt PN codethat reflects the current estimated code phase; the late output providesa late PN code that is delayed in time with respect to the prompt PNcode by a known time period (e.g., one chip). If correlations areavailable between the received signal and the early, prompt and latevariants of the locally generated replica of the received signal, thereceiver 11 may adjust the phase and time delay (e.g., via shiftregisters) of the locally generated replica in an attempt to maximizecorrelation, for example.

In one embodiment, the first signal generator 32 may comprise anygenerator for generating a spread spectrum code, spread spectrumsequence, binary sequences, Gold codes, PN code, a pseudo-random noisecode sequence, or a PN code that is similar to a spread spectrum code,spread spectrum sequence, binary sequences, Gold codes, pseudo-randomnoise code, pseudo-random noise code sequence, or a PN code transmittedby a transmitter of a satellite for reception by the receiver 11 as thecomposite received signal. In another embodiment, the first signalgenerator 32 may be formed of series of shift registers that are loadedwith an initial starting code sequence, where the shift registers havevarious selectable or controllable taps for providing feedback andreiterative values as the output.

A second signal generator 31 (e.g., BOC/QBOC generator) generates alocal replica of a BOC and QBOC waveform. The first multiplexer 30 andthe second multiplexer 29 may be referred to as the first selector andthe second selector. The first selector (30) selects a time period orchip, in terms of chip phase, either advancing, synchronizing, ordelaying the received sample (101 or 102); the second selector (29)picks either the BOC or QBOC waveform, whose output is mixed with theselected PN chip sequence to generate either BOC-PN replica or QBOC-PNreplica for received sample 101.

The first signal generator 32 and the second signal generator 31provides output signals to one or more correlator modules 130 (e.g. codecorrelators). The first multiplexer 30 and the second multiplexer 29each have a select input (123, 124) to select which mux input terminalis routed to a mux output terminal for the multiplexers (29, 30), wherethe select input can be determined by the decision unit 35 as explainedin detail later. The mux output terminals are fed into the multiplier orcode mixer 42 to create a locally generated replica of either BOC-PN orQBOC-PN signal for correlation to the received composite sample (101 or102). The output of the multiplexer or code mixer 42 is provided to thecode wipe-off module 27 to remove BOC-PN or QBOC-PN modulation from thereceived signal (101 or 102). The three versions of PN code by the firstsignal generator 32 can interact with the two versions of squaredwaveform BOC or QBOC from the second signal generator 31 to producevarious permutations of local replica signals to generate differentcorrelations through the integration-and-dump module 28. A bank ofcorrelations 159 is used for decoding, demodulating, CD and CR phasetracking.

Here, the code wipe-off module 27, together with integration-and-dumpmodule 28, produces correlations or correlation values between thereceived signal and the locally generated BOC-PN component to generateBOC I vector data, BOC Q vector data; the code wipe-off module 27produces correlations or correlation values between the received signaland the locally generated QBOC-PN component to generate QBOC I vectordata, QBOC Q vector data. In one configuration, the BOC correlations areassociated with a BOC correlation function, whereas the QBOCcorrelations are associated with a QBOC correlation function.

A decision unit 35 or electronic data processor decides whether: (a) togenerate the BOC I vector data and BOC Q vector data for any given timeinterval or sampling period, where the BOC correlation function is usedto demodulate the received signal for the sampling period; or (b) togenerate the QBOC I vector data and the QBOC Q vector data for any giventime interval or sampling period, where the QBOC correlation function isused to demodulate the received signal for the sampling period. A firstdetector 201 (e.g., first signal envelope/amplitude detector) and asecond detector 211 (e.g., second signal envelope/amplitude detector)provide primary amplitude and secondary amplitude data for the receivedI and Q vector data to the decision unit 35 or data processor.

At the antenna 20, the received composite signal can have the totalsignal power or total signal energy divided between the BOC component orQBOC component of the signal, where a greater primary amplitude than asecond amplitude indicates that a majority of the total signal power isin the BOC component during any sampling interval (e.g., samplingepoch); a greater secondary amplitude than primary amplitude indicatesthat a majority of total signal power is in the QBOC component duringany sampling interval. The data processor or decision unit 35 selects(e.g., via the first and second multiplexers 29, 30) a set of BOCcorrelations in accordance with a BOC correlation function for thesampling period if the primary amplitude exceeds the secondary amplitudefor the sampling period. The data processor or decision unit 35 selectsa set of QBOC correlations in accordance with a QBOC correlationfunction for the sampling period if the secondary amplitude exceeds theprimary amplitude for the sampling period.

In general, the data processor or correlator module 130, at everysampling epoch or sampling period, uses either BOC correlation functionor the QBOC correlation function, whichever retains more energy, tosupport unambiguous CD pull in of the received signal by eliminating thefalse zero-crossing points of the discriminator S-curve. Suchamplitude-driven selection is named as decision-directed selection(DDSel). An epoch means a specific instant in time of a navigationsatellite system or the time interval during which the receiver 11measures the carrier phase at a corresponding frequency or rate. In thedigital receiver portion 192, the data processor, at every samplingepoch, can also use multiple BOC correlation with different chip spacingto eliminating the false zero-crossing points of its discriminatorS-curve. One or more false maxima or local maxima of a correlationfunction may result in the false zero-crossing points on thediscriminator S-curve. If more than one zero-crossing point is presentin the discriminator S-curve, the zero-crossing points can presentambiguity to detecting the maximum correlation.

In one embodiment, in the digital receiver portion 192 the dataprocessor forms the correlation function by selecting the greateramplitude, either the primary amplitude derived from BOC component orthe secondary amplitude derived from QBOC component. The resultant DDSelamplitude estimation, FIG. 9, eliminates all the zero-crossing points,within +/−1 chips, being present on either BOC or QBOC correlationfunctions as shown by FIG. 8.

The decision unit 35 selects either the BOC correlations (including BOCI vector data, BOC Q vector data) or QBOC correlations (including, QBOCI vector data and QBOC Q vector data) for use in the positionmeasurement, position estimation, or attitude estimation (e.g., tilt,roll and yaw angle) and for tracking of the code and carrier of thereceived composite signal. For example, the decision unit 35 controlsthe first multiplexer 30 (e.g., first selector), the second multiplexer(e.g., second selector) or both to form BOC correlations, QBOCcorrelations, or both that are aligned with or track the compositereceived signal for each carrier or channel.

The CR tracking module 38 and CD tracking module 37 are collectivelyreferred to as the tracking module 200 in FIG. 1 and FIG. 2. Thetracking module 200 supports measurement of the CR frequency, CR phase,and CD phase (e.g., individually or collectively, measurement data) ofthe received signals to control one or more locally generated referencesignals with respect to corresponding received signals (derived from thecomposite received signal) to maximize correlation of the correspondingreceived signals to the respective locally generated reference signals.In one embodiment, there are multiple received signals because thereceiver 11 receives four received signals from at least four differentsatellite transmitters to estimate the position of the receiver antenna20. For example, the CD tracking module 37 or CR tracking module 38 cangenerate measurement data that the data processor or receiver uses tocontrol an adjustable time delay (e.g., route data through known numberor sequence of shift registers), or engage in other data processing ofone or more digital signals associated with a locally generatedreference signal with respect to the received signal to maximizecorrelation of each received signal to the corresponding locallygenerated reference signal.

In one embodiment, the carrier tracking module 38 may comprise one ormore of the following: a CR measurement module, a CR phase counter, afrequency detector, and a phase detector. For example, the CR phasecounter counts both the number of integer cycles plus the fractionalcycles of the received CR during a known time period and a phasedetector that measures the fractional CR phase of the received CR at aninstantaneous time or sampling interval within the known time period tosynchronize the receiver's CR phase with the observed or measured CRphase of the received signal.

In certain embodiments, the code tracking module 37 is arranged forgenerating a control signal, a clock signal, or shifting an adjustabletime delay of the locally generated reference CD signal with respect tothe digital received signal in response based on maximizing correlationof the received signal to the locally generated reference code signal.

In one embodiment, the tracking module 200 and corresponding NCO's (33,34) are collectively adapted to shift an adjustable time delay of one ormore locally generated reference signals with respect to the digitalreceived composite signal (or its CD and CR components) in response tocontrol data, measurement data, or both provided by the tracking loopsignal processor of the tracking module 200 based on maximizingcorrelation of the received composite signal (or its code and carriercomponents) to the locally generated reference signal.

A carrier tracking module 38 facilitates alignment of the phase of thelocally generated replica of the CR to the received signal. The carriertracking module 38 provides control data or a control signal to acarrier numerically controlled oscillator (NCO) module 34 to adjust thelocally generated replica signal of the CR produced by carrier NCOmodule 34. In one embodiment, the carrier NCO module 34 provides thelocally generated replica of the carrier to the carrier wipe-off module26. The carrier NCO module 34 may receive an input reference clocksignal and output an adjusted clock signal or another control signal forgenerating the locally generated CR frequency that accurately alignswith the CR phase or the residual carrier phase of received sample (101or 102).

The code tracking module 37 facilitates alignment of the phase of thelocally generated BOC-PN or QBOC-PN replica with respect to the receivedsample (101 or 102). The CD tracking module 37 provides control data ora control signal to adjust a code numerically controlled oscillator(NCO) module 33, where the code NCO module 33 controls the chipping rateof the first signal generator 32 and a binary sine or cosine waveformusing the second signal generator 31. The code tracking module 37, whichnormally comprises a delay locked loop (DLL), generates a control signalto tune the chipping rate of code NCO module 33. The CD phase, output ofcode NCO module 33, is used to drive the first signal generator 32(e.g., PN sequence generator) and the second signal generator 31 (e.g.,BOC/QBOC generator). Multiple local PN sequences are generated by thefirst signal generator 32; the local PN waveform either advances,synchronizes, or delays its phase against the CD phase of the receivedsample 101.

The code mixer 42 mixes or multiplies the output of the firstmultiplexer 30 and the second multiplexer 29. The first output of thefirst multiplexer 30 may comprise an early PN code, a prompt PN code, ora late PN code. The second output of the second multiplexer 29 maycomprise a locally generated BOC signal or a QBOC signal. The firstmultiplexer 30 has first inputs that are coupled to output of an earlyPN code, a prompt PN code, or a late PN code. The second multiplexer 29has second inputs that are coupled to a BOC generator, a QBOC generator,or the combination of a BOC/QBOC generator.

The data demodulator 40 provides satellite navigation data forestimating a range (e.g., distance between a satellite and the antenna20) or a position (e.g., in two or three dimensional coordinates) ofphase center of the antenna 20. The satellite navigation data or othersignal information may comprise one or more of the following informationthat modulates the baseband waveform of the received signal: date,satellite navigation system time, satellite status, orbital data,ephemeris data, almanac, satellite location, and satellite identifier.The data demodulator may use phase shift keying, phase demodulation,pulse width demodulation, amplitude demodulation, quadrature amplitudedemodulation, or other demodulation technique that is consistent withthe modulation by the modulator at the satellite transmitter.

In one embodiment, the data demodulator 40 outputs a demodulated signalor demodulated encoded data, such as a demodulated digital signal with aquadrature phase component and in-phase component at baseband. The datamay comprise one or more following information such as date, satellitenavigation system time, satellite status, orbital data, ephemeris data,almanac, satellite location, and satellite identifier.

In one embodiment, the measurement generation module 39 measures thepropagation time between transmission of a satellite signal from acertain satellite to the receiver antenna 20 and converts thepropagation time into a distance or range proportional to the speed oflight. The measurement generation module 39 determines a range,pseudo-range or estimated range between the receiver antenna 20 and fouror more satellites with a reliable signal quality or signal strengthbased upon one or more of the following: (a) the measured CD phase ofeach received signal, and (b) the measured CR phase of each receivedsignal. In one embodiment, the measurement generation module 39 mayresolve ambiguities in the measured CR phase of the received signal bysearching for a solution that is consistent with one or more of thefollowing: (1) a position estimated from decoding the code portion ofthe signal, (2) known reference position of the receiver antenna 20, and(3) differential correction data applicable to the received signal.Further, the measurement generation module 39 may be associated with awireless receiver (e.g., satellite receiver, mobile transceiver, orcellular transceiver) that receives navigation correction data from areference satellite navigation receiver to reduce or eliminate sourcesof bias or error (e.g., certain clock errors or propagation errors) inthe CR phase measurements.

The navigation positioning engine 41 determines the position estimate ofthe receiver antenna 20 based on the measured CR phases, estimatedranges of the measurement generation module 39 and demodulated data. Forexample, the positioning engine 41 may use ranges from four or moresatellites to determine the position, velocity, or acceleration of theantenna 20 of the receiver in two or three dimensions.

In the digital receiver portion 192, the receiver 11 or its dataprocessing system may comprise hardware and software instructions. Forexample, in one illustrative embodiment the hardware comprises a dataprocessor that communicates to a data storage device, which storessoftware instructions, via one or more data buses.

In the digital receiver portion 192, as used throughout the document thedata processor may comprise one or more of the following: an electronicdata processor, a microprocessor, a microcontroller, an applicationspecific integrated circuit (ASIC), digital signal processor (DSP), aprogrammable logic device, an arithmetic logic unit, or anotherelectronic data processing device. In the digital receiver portion 192,the data storage device may comprise electronic memory, registers, shiftregisters, volatile electronic memory, a magnetic storage device, anoptical storage device, or any other device for storing data. The dataprocessor may be coupled to the data storage device via one or more databuses, which support communication between the data processor and thedata storage device. As used herein the, data processor may refer to oneor more components or modules of the digital receiver portion 192,including by not limited to any of the following: the carrier wipeoffmodule 26, the code wipeoff module 27, the integration-and-dump module28, correlator module 130, mixer 42, second multiplexer 29, firstmultiplexer 30, first signal generator 32, second signal generator 31,first detector 201, second detector 211, decision unit 35, accumulatormultiplexer 36, code tracking module 37, carrier tracking module 38,measurement generation module 39, data demodulator 40, and navigationpositioning engine 41.

In general, the digital receiver portion 192 comprises computer or anelectronic data processing system that comprises an electronic dataprocessor, digital logic circuits, multiplexers, multipliers, digitalfilters, integrators, delay circuits, oscillator, signal generator, PNcode sequence generator, registers, shift registers, logic gates, orother hardware. The electronic data processing system may supportstorage, retrieval and execution of software instructions stored in adata storage device.

In one embodiment, the digital receiver portion 192 or electronic dataprocessing system is capable of extracting a BOC component from thecomposite received signal by combining the received composite signalwith a local BOC-PN replica (e.g., at the output node of code mixer 42).The digital receiver portion 192 is capable of deriving a QBOC componentby combining with a local QBOC-PN replica (e.g., at the output node ofthe code mixer 42). The BOC component comprises an in-phase BOCcomponent and a quadrature-phase BOC component, the QBOC componentcomprising in-phase QBOC component and a quadrature-phase QBOCcomponent. The digital receiver portion 192 advantageously decideswhether to select the local BOC-PN replica or the local QBOC-PN replicato generate the CD and CR control signal to CD tracking module 37 and CRtracking module 38. Such selection minimizes false locks that mightotherwise occur with respect to a correlation function associated solelywith a BOC early minus late (EML) error function or a QBOC correlationEML function. For example, a BOC EML error function comprises an earlyBOC correlation and corresponding late BOC correlation for a samesampling period or epoch; QBOC EML function comprises an early QBOCcorrelation and corresponding QBOC correlation for a common samplingperiod or epoch.

FIG. 2A and FIG. 2B are collectively referred to as FIG. 2. FIG. 2illustrates one possible realization of the digital receiver portion 192or digital processing section of the receiver 11 in greater detail thanFIG. 1. Like reference numbers in FIG. 1 and FIG. 2 indicate likeelements.

FIG. 2 illustrates an example of the realization of one or morecorrelators (130) of a channel using a window correlation technique(e.g., WinCorr). Here, the WinCorr technique provides a flexibleresolution to observe any section (e.g., fraction or window) of a chip.As illustrated, the digital received sample 101, with residualfrequency, from the ADC 24 is inputted to the CR wipe-off module 26 at asuitable sampling rate (e.g., at least in the MHZ range). The CRwipe-off module 26 mixes the digital received sample with local CRreplica 103. For example, the CR wipe-off module 26 mixes the digitalreceived sample with a synchronized CR carrier replica 103 through adigital CR phase rotator to wipe off the carrier. The output of the CRwipe-off module 26 is inputted to a set of correlator modules 130, suchthat each received channel (e.g., each modulated carrier) is assigned(dynamically or statically) at least one correlator module 130. Eachcorrelator module 130 may comprise at least a code wipe-off module 27and an integration-and-dump module 28.

In one embodiment, the set of the correlator modules 130 despreads thePN code and generates a group of accumulation products or correlationdata 125 at millisecond or multi millisecond rates for CR tracking andCD tracking by the tracking module 200. The correlation data 125 (e.g.,correlations or accumulation products) are stored in a data storagedevice associated with the digital receiver portion 192 of the receiver11. Although FIG. 2 shows three correlator modules 130 as dashedrectangular blocks, any number of correlator modules 130 can be used.For example, a BOC group of correlator modules 130, a QBOC group ofcorrelator modules 130, or both can be used for each received channel toproduce a corresponding early, late or prompt correlation oraccumulation for each received channel, where illustrative possiblepermutations of the correlation data 125 are within correlation datablock 125 in FIG. 2. The correlation data 125 can be generallycharacterized in accordance with the representation of xBOC(TAP,WinX),where xBOC specifies a BOC or QBOC correlation, WinX specifies an early(E), prompt (P) or late (L) correlation, and TAP indicates a particulartap or output of a respective correlator module 130.

The first signal generator 32 and the second signal generator 31 provideinput data to one or more correlator modules 130. The first signalgenerator 32 (e.g., PN coder) provides the early (E), punctual or prompt(P), and late (L) locally generated replicas of the code signals (118,117, and 116, respectively) for the BOC EML function, the QBOC early EMLfunction, or both. The second signal generator 31 (e.g., BOC/QBOCgenerator) provides a BOC signal 114 and a QBOC signal 115. A first codemixer 141 modulates BOC waveform on the selected chip 119 to form aBOC-PN signal 140 (e.g., BOC(TAP)). The second code mixer 121 modulatesQBOC waveform on the selected chip 119 to form corresponding QBOC-PNsignal 120 (e.g., QBOC(TAP)). The first signal generator 32 provides thePN code signals (early, prompt and late variants of the PN code signal)to a multiplexer 30. In turn, the multiplexer provides one of PN codedsignals to the inputs of the first code mixer 141 and the second codemixer 121. The first signal generator 32 and the second signal generator31 are synchronously controlled by the CD NCO module 33. An individualwindow control signal or data comprises a lower window limit 110 and anupper window limit 111 that is provided to the code phase window module157. Each correlator module 130 of the channel may have different windowconfiguration.

In FIG. 2 (collectively FIG. 2A and FIG. 2B), through tap selectionsignal 124, the PN-BOC signal 140 or the PN-QBOC signal 120, in terms ofcode phase, can advance, synchronize, or delay against the CD phase ofreceived sample (101 or 102). The second multiplexer 29 (or selector) isused to select the local BOC-PN signal 140 or the local QBOC-PN signal120 as locally generated replica signal 105 at the output of logicmodule 106 for generation of BOC or QBOC correlations for any samplingperiod or duration. In one particular configuration, the correlationsmay be consistent with the BOC early-minus-late error (EML) function orthe QBOC early-minus-late (EML) error function, for any sampling periodor duration. The SEL_BOC signal or selection signal 123 determineswhether to select the BOC or QBOC correlation functions based onamplitude measurements of the primary amplitude and secondary amplitude.

The code phase window module 157 provides the flexible resolution orvariable window size to visualize any section of the chip of the PNcode. For example, the fractional chip phase signal 109 compares withthe two window limits, the lower limit 110 (e.g., Win(k):x1) and theupper limit 111 (e.g., Win(k):x2). If the chip phase is located withinobserving section, the code wipe off module (27) is enabled to operateand to provide a non-zero output for input to integration-and-dumpmodule 28 for accumulating, otherwise (if the chip phase is locatedoutside of the observing section) zero output is fed into the module 28for integrating. During window enable period, the local signal 108(e.g., xBOC(TAP)) is used to despread the PN modulation of the receivedsample (101 or 102). The correlator modules 130 despread the localsignals 108 that may comprise, but are not limited to, various availablesignal versions or signal permutations. Here, the available signalversions for the local signals 108 are named E-BOC, E-QBOC, P-BOC,P-QBOC, L-BOC, L-QBOC, where E means early, P means prompt and L meanslate, and where Q means quadrature. The output of the code phase windowmodule 157 and the second multiplexer 29 are applied to a logic module106. In one embodiment, the logic module 106 may comprise an AND gate orother suitable logic device, logical circuit or logic function. Theoutput of the logic module 106 is coupled to code wipe off module 27 toprovide an enable signal and window limit for accumulating thedespreaded samples by integration-and-dump modules 28.

Multiple correlator modules 130 provide a bank of correlation data 125for the tracking module 200. The code wipe off module 27 convolutes thelocal replica signal (105 or 108) with the baseband signal 102 todespread the BOC and PN modulation. The integration-and-dump module 28integrates multiple dispreading samples over millisecond or multiplemillisecond periods. The integrated output correlation data 125 frommultiple correlators 130 may comprise one or more of the followinginformation in encoded or modulated form: date, satellite navigationsystem time, satellite status, orbital data, ephemeris data, almanac,satellite location, and satellite identifier, for example. Eachcorrelator module 130 comprises an integration- and dump-module 28 thatgenerates a corresponding correlation or accumulation. The correlationsor accumulations are stored or held in a data storage device, such aselectronic memory, a register, or another device within the digitalreceiver portion (e.g., 192) for subsequent or real-time processing bythe decision unit 35 and the tracking module 200, among other things.

Depending on the tap selection signal 124, BOC selection signal 123, andwindow limit data (110, 111) or control signals associated with windowlimit data (110, 111), each correlator module 130 is configurable togenerate one of various correlation data as but not limited to thoselisted in correlation data 125. Window control signals, associated withwindow limit data 110 and 111, provide different resolution to a chip ofthe PN code which determines the pull in range and the lock accuracy.Normally, observing a small section of the chip by the code phase windowmodule 157 improves the tracking accuracy while minimizing the pull-inrange; while an observing a large section of the chip extends thepull-in range at the expense of accuracy. For example, the tap selectionsignal 124 (e.g., SEL_TAP) provides a flexible visualization acrossmultiple chips to examine the correlation between the received samples(101 or 102) and its next chip by selecting E tap for its correspondingearly chip, by selection P tap for its corresponding prompt chip, byselecting L tap for its corresponding late chip. The BOC selectionsignal 123 provides an efficient way to recover the energy from thereceived sample (101 or 102) with BOC modulation by either extractingits BOC correlation component or deriving its QBOC correlationcomponents.

In FIG. 2, a set of correlation data 125 is stored in a register or setof registers, a processor or data storage device for storingintermediate results or other data. In one embodiment, correlation data125 comprise correlations, correlation values, or outputs of one or morecorrelators of the correlator module 130 for processing to support theformation of or realization of a dual early-minus-late error functionfor each BOC channel and each QBOC channel, for example.

For each channel, the correlation data 125 provides essentialinformation to extract the information such as amplitude, CD phasemisalignment and CR phase misalignment from either BOC or QBOCcorrelation output. As shown by FIG. 8, depending on the CD phasealignment, the recovered energy to input sample (101 or 102) eitherlocates at BOC component or QBOC component. The multiple local maximapresented on FIG. 8 possibly deteriorate the CD tracking by incurringfalse lock point on the discriminator S-curve. The multiple zero pointspresented on FIG. 8 negatively impact the amplitude estimation which isimportant to correctly estimate CD and CR tracking error through anormalization process; for example, the zero BOC correlation componentmisleads to a decision of a false signal loss (or perceived false cycleslip) while the energy still exists on the QBOC correlation component,and vice versa. Therefore, the strategy named DDSel selects either theset of BOC correlations or the set of QBOC correlations at every epoch(or sampling period) to facilitate amplitude estimation, CR tracking andCD tracking. DDSel amplitude estimation, compared to using only BOC orQBOC demodulated component, takes advantage by always choosing thecomponent with most of demodulation energy to minimize noise infection;DDSel CD tracking not only eliminates the false zero-crossing points ondiscriminator S-curve, but also reflects improvement in terms of bothreliability and accuracy. DDSel CR tracking also reveals the improvementin both reliability and accuracy.

The output of every correlator within the correlator module 130 containsan I part and a Q part. Multiple correlations, not limited to thoselisted as correlation data 125, support the envelope detector 300(reflected by detectors 201 and 211 at FIG. 3), CD tracking module 37,and CR tracking module 38. As used herein, receiver channel informationmeans one or more of the following data: (a) code phase 109 from code(CD) numerical control oscillator (NCO) 126, (b) late (L) tap 116,punctual (P) tap 117, or early (E) tap 118 from a first signal generator32 (e.g., PN Coder), (c) BOC signal 114, QBOC signal 115 from a secondsignal generator 31 (e.g., BOC/QBOC generator), and (d) local carrier(CR) replica 103 (e.g., in sine or cosine form) resulting from carrierNCO module 34.

The receiver channel information is fed into the set of the correlatormodules 130. In one configuration, each correlator module 130 functionsor operates in accordance with one or more of the following parameters:(a) a lower window limit 110 (e.g., signal or data) in chips (e.g., orfractional chips), (b) an upper window limit 111 (e.g., signal or data)in chips, (c) BOC versus QBOC selection via selection BOC selection data123, and (d) tap selection signal 124 for selection of the early (E),prompt (P) or late (L) versions or variants of the PN code forapplication to the first and second code mixer (141, 121). The output ofeach correlator of the correlator module 130 varies based on the outputof the first code mixer 141 (e.g., tap BOC code mixer), and second codemixer 121 (e.g., tap QBOC code mixer).

In one configuration, each correlator module 130 comprises a codewipe-off module (e.g., 27 in FIG. 1A) that supports wiping off (e.g.,removing) the PN modulation using a locally generated replica signal 105(e.g., XBOC(TAP, Win). The set of accumulations or correlation data 125,but not limited to those listed in FIG. 2 or FIG. 3, are processed bythe carrier tracking module 38 and code tracking module 37. Theresultant code rate 260 (e.g., code NCO rate) and carrier rate 270(e.g., carrier NCO rate) are used to drive the code NCO module 33 andcarrier NCO module 34 in FIG. 2. In one embodiment, the code trackingmodule 37 comprises a code error module 43 linked or coupled to a codeloop module 44. In one embodiment, the carrier tracking module 38comprises an Automatic Frequency Control (Costas Error) module 45 linkedor coupled to the carrier loop module 46.

FIG. 3A and FIG. 3B collectively may be referred to as FIG. 3. FIG. 3illustrates the tracking module 200 of FIG. 2 in greater detail thanFIG. 2. Further, FIG. 3 illustrates the envelope detector (201, 211) anddecision unit 35.

In FIG. 3, a first detector 201 (e.g., first envelope detector) detectsa primary amplitude of the BOC component. The first detector 201 maycomprise an envelope detector that detects the signal energy or power ofthe BOC component, including an in-phase (I) component of BOCcorrelation and a quadrature (Q) component of BOC correlation. Forexample, the first detector 201 may detect the primary amplitude as theaggregate power of the I vector of the prompt BOC component and the Qvector of the prompt BOC component.

A second detector 211 detects a secondary amplitude of the QBOCcomponent. The second detector 211 may comprise an envelope detectorthat detects the signal energy or power of the QBOC component comprisingin-phase QBOC component and a quadrature-phase QBOC component. Forexample, the second detector 211 may detect the secondary amplitude asthe aggregate power of the I vector of the prompt QBOC component and theQ vector of the prompt QBOC component.

The first detector 201 and the second detector 211 provide output of theprimary amplitude and secondary amplitude or energy data on the receivedsignals to a data processor, an evaluator, or the decision unit 35. Thedecision unit 35 comprises an electronic data processor or evaluator.The decision unit 35 provides the amplitude estimation by selectingeither the primary amplitude or the secondary amplitude, whichever isgreater; it also outputs a logic signal 210 (e.g., at a correspondingnode indicated in FIG. 2) to indicate whether the primary amplitudeexceeds the secondary amplitude. If the channel operates at steady statemode, the selection signal (SEL_BOC) 123 always selects the set of BOCcorrelations to drive the CD and CR tracking loop; otherwise, theselection signal (SEL_BOC) 123 synchronizes with the signal 210. Theselection signal 123 determines whether to use BOC component or QBOCcomponent to estimate the CD and CR error, such as the multiplexor 237for CD error, the multiplexor 238 for CR frequency error, and themultiplexor 239 for CR phase error.

In FIG. 3, the decision unit 35 generate two signals; the amplitudeestimation signal 273 reflects the power/amplitude of demodulatedsignal; the logic signal 210 has a first state (e.g., true) if theprimary amplitude from BOC component is greater than the secondaryamplitude from QBOC component, otherwise the logic signal 210 has asecond state (e.g., false) that is different than the first state.

The steady state operation mode of the receiver 11 occurs after pull-inor lock-on of the code phase of the received signal. In steady stateoperation mode for any received composite signal, the local CD and CRphase estimation tightly synchronizes with the phase (e.g., of the CDand CR) of the received sample (101 or 102) in FIG. 2; therefore, thedemodulation energy is concentrated in the BOC component while leavingQBOC component dominated by noise. To mitigate the negative impact ofnoise, if using DDSel method during the pull-in mode, only the set ofBOC correlation data 125 is preferred at steady state. After codeacquisition (e.g., or pull-in) of the received signal and during asteady state mode in which a local code replica and a local carrierreplica is synchronized with a phase, or respective code and carrier, ofthe received signal over one or more sampling periods that meet orexceed a threshold duration, the digital receiver portion 192 or dataprocessor can use only the set of BOC correlations to drive the code andcarrier tracking.

In an alternate embodiment, during the pull-in mode the receiver (e.g.,11) may use multiple BOC correlations with different windows (referredto as MWin-BOC), instead of the DDSel method during the pull-in mode. Ifthe receiver uses multiple BOC correlation with different windows(MWin-BOC) method during the pull-in mode, then for the steady statemode the correlation with a single window (referred to as SWin-BOC) ispreferred to formulate the CD error estimation.

The receiver or data processor determines whether the operation modeshould be the pull-in mode or the steady state mode by analyzing theprimary amplitude. In one embodiment, if the detector or receiverdetermines that the primary amplitude is continuously greater than thesecondary amplitude for equal to or greater than a threshold duration(e.g., number of epochs or sampling intervals), the decisionaugmentation module or counter 220 drives the CD tracking module 37 andCR tracking module 38 of FIG. 2 into a steady state mode. In oneembodiment of the steady state mode, the receiver (e.g., 11) assumes thecompletion of CD and CR pull-in process (either using DDSel method orMWin-BOC method) and only uses the BOC correlations to drive the CR andCD tracking. If the signal window-BOC signal 242 (e.g., SWin-BOC) isenabled, only the set of BOC correlations contributes to the CD and CRtracking operation. In another embodiment, if the detector or receiverdetermines that the primary amplitude is continuously greater than orequal to a threshold amplitude for greater than a threshold duration,the decision augmentation module or counter 220 drives the CD trackingmodule 37 and CR tracking module 38 of FIG. 2 into a steady state mode.

FIG. 3 illustrates two CD tracking methods operating at pull in stage,the selection signal 234 (SEL_MWIN_BOC) determines either to use DDSelCD tracking method or MWin-BOC method. DDSel CD tracking methodgenerates the CD error estimation using BOC or QBOC correlations,whichever contains most of the energy. The MWin-BOC method combinesmultiple BOC correlations with different chip spacing to estimate the CDerror. Both methods achieve the same purpose, eliminating the falsezero-crossing points on discriminator S-curve.

FIG. 3 addresses the DDSel estimations for CD error 231, CR frequencyerror 235 (e.g., fDDsel), and CR phase error 236, which minimize theimpact of the noise disturbance by selecting either the set of BOCcorrelation or the set of QBOC correlation, whichever contains most ofthe demodulated energy. As shown in FIG. 3, the CD discriminators 202,212 and 203 are noncoherent, i.e., using both the I vectors and Qvectors to estimate CD error, such method solves or determines thepreviously unknown allocation of energy between the I component and Qcomponent.

FIG. 3 illustrates the signal processing chain to explain the details ofdetermining DDSel and MWin-BOC technique. A mathematical model for BOCand QBOC using window correlation (e.g., WinCorr) techniques isintroduced. As mentioned before, a local QBOC retains the missing partof signal energy if the local BOC replica signal hasn't perfectlyaligned with the received BOC signal. The mathematic model of BOC andQBOC component to a general window correlation configuration (TAP,W_(X)) are modeled as the following equations:

$\begin{matrix}\begin{matrix}{{TAP}_{BOC}^{W_{X}} = {{AW}_{X}\sum_{{i = 0},{{di} = {F_{Chip}/F_{S}}}}^{N_{Chip}}}} \\{\left\{ \begin{matrix}\begin{matrix}{c_{i}{c_{i - \tau} \cdot}} \\{{\sin\left( {2\pi\; N_{BOC}i} \right)}{{\sin\left( {2\pi\;{N_{BOC}\left( {i - \tau} \right)}} \right)} \cdot}}\end{matrix} \\{\mathbb{e}}^{j\;{\delta\theta}_{i}}\end{matrix} \right.} \\{= {\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}{\mathbb{e}}^{j\;\delta\;\varphi}}} \\{= {\overset{\overset{I\;\_\;{TAP}_{BOC}^{W_{X}}}{︷}}{\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}\cos\;\delta\;\varphi} +}} \\{\underset{\underset{Q\;\_\;{TAP}_{BOC}^{W_{X}}}{︸}}{j\;\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}\sin\;\delta\;\varphi}}\end{matrix} & {{Equation}\mspace{14mu} 1} \\\begin{matrix}{{TAP}_{QBOC}^{W_{X}} = {{AW}_{X}\sum_{{i = 0},{{di} = {F_{Chip}/F_{S}}}}^{N_{Chip}}}} \\{\left\{ \begin{matrix}\begin{matrix}{c_{i}{c_{i - \tau} \cdot}} \\{{\sin\left( {2\pi\; N_{BOC}i} \right)}{{\sin\left( {2\pi\;{N_{BOC}\left( {i - \tau} \right)}} \right)} \cdot}}\end{matrix} \\{\mathbb{e}}^{j\;{\delta\theta}_{i}}\end{matrix} \right.} \\{= {\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}{\mathbb{e}}^{j\;\delta\;\varphi}}} \\{= {\overset{\overset{I\;\_\;{TAP}_{BOC}^{W_{X}}}{︷}}{\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}\cos\;\delta\;\varphi} +}} \\{\underset{\underset{Q\;\_\;{TAP}_{QBOC}^{W_{X}}}{︸}}{j\;\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}\sin\;\delta\;\varphi},}\end{matrix} & {{Equation}\mspace{14mu} 2}\end{matrix}$where

-   -   TAP_(BOC) ^(w) ^(X) is the correlation based on the BOC input        and BOC local replica with a selection of PN TAP, TAP can be        early (E), prompt (P), or late (L),    -   TAP_(QBOC) ^(w) ^(X) is the correlation based on the BOC input        and QBOC local replica with a selection of PN TAP,    -   i is chip phase in unit of chips,    -   N_(Chip) is number of chips per millisecond,    -   di is sample phase increment in unit of chips,    -   A is the amplitude of signal at level of millisecond        integration,    -   W_(X) is the window size in chips,    -   N_(BOC) is equal to m/n of BOC (m,n), where m is f_(m)/f_(c) and        n is f_(n)/f_(c), f_(m) is a first subcarrier frequency, f_(n)        is the actual chip frequency, and f_(c) is the reference        chipping rate,    -   F_(Chip) is the chipping rate in chips/second,    -   F_(S) is the sampling rate in samples/second,    -   R(τ) is the conventional correlation function for PN code,    -   τ is the chip phase error estimated at receiver side,    -   δθ_(i) is the carrier phase estimation error for each sample,    -   δφ is the average carrier phase estimation error over N_(Chip)        period,    -   I_TAP_(BOC) ^(w) ^(X) is the in-phase or real part of TAP_(BOC)        ^(w) ^(X) ,    -   Q_TAP_(BOC) ^(w) ^(X) is the quadrature or imaginary part of        TAP_(BOC) ^(w) ^(X) ,    -   I_TAP_(QBOC) is the in-phase or real part of TAP_(QBOC) ^(w)        ^(X) ,    -   Q_TAP_(QBOC) ^(w) ^(X) is the quadrature or imaginary part of        TAP_(QBOC) ^(w) ^(X) .        Equation 1 and Equation 2 show that cos(2πN_(BOC)τ) and        sin(2πN_(BOC)τ) are orthogonal against τ, i.e., as shown in FIG.        8, P_(BOC) ^((0,1)) (indicated by a solid curved line 701) and        P_(BOC) ^((0,1))(indicated by a dashed curved line 702), the        amplitude (correlation output) of the curved lines (701, 702)        reach or cross zero at alternating times measured in chips.        Curved line 701 represents the BOC correlation function of        Equation 1, whereas curved line 702 represents the QBOC        correlation function of Equation 2. The horizontal axis 704 of        FIG. 7 shows a time difference between the received signal and        locally generated replica signal in chips, whereas the vertical        axis 703 shows correlation between the received signal and        locally generated replica signal.

Equation 1 and Equation 2 approximates the binary sub-carrier by using asine or cosine model to remove the discontinuity of sign( ) functionwhich reflects the binary characteristics. The model of Equation 1 andEquation 2, through extensive comparison, effectively reflect thecharacteristics of BOC and QBOC correlations with binary characteristicsincluded. For example, the sign( ) function typically outputs a firstlogic level or a positive input and a second logic level, different thanthe first logic level, for a negative input.

In FIG. 3, in one embodiment the decision unit 35 makes the BOC/QBOCselection resulting from the DDSel technique. The decision unit 35 doesone or more of the following: (a) compares the primary amplitude withthe secondary amplitude, and selects the greater one to update theamplitude estimation; sets signal 210 with true (e.g., or a first logicstate) if the primary amplitude exceeds the secondary amplitude, orreset it otherwise (e.g., or set with second logic state opposite to thefirst logic state), (b) provides the DDSel data for control of theselection multiplexers (237, 238, 239) to choose either BOC CD and CRerrors or QBOC CD and QBOC CR errors for tracking in the CD loop and CRloop, and adjustment of the CD and CR phase to maximize correlationbetween the received signal and locally generated replicas.

If the primary amplitude continuously exceeds the secondary amplitudefor more than M epoch, the most demodulation energy concentrates in theBOC component, i.e., the local BOC-PN replica assumedly aligns with theCD phase of input sample 101 (FIG. 2). Under such circumstance, thechannel could operate at steady state mode which set signal 242, thusturning on the BOC selection signal 123.

The first detector 201 and the second detector 211 may use Equation 3 orEquation 4 to determine or estimate the amplitude of the BOC or QBOCsignal components.Amp(Y)=√{square root over (I_Y ² +Q_Y ²)}, where  Equation 3:

I_Y is the in-phase xBOC component,

Q_Y is the quadrature-phase xBOC component, and

xBOC refers to the BOC or QBOC signal component of the receivedcomposite signal. Under above Equation 3, the primary amplitudecomprises the combined signal power of the BOC in-phase component andBOC quadrature phase component. Similarly, the secondary amplitudecomprises the combined signal power of the QBOC in-phase component and aQBOC quadrature-phase component. Equation 3 is ideal amplitudecalculation based on in-phase and quadrature components. However alinear approximation for Equation 3 is as follows:Amp^(approx)(Y)

max(|I_Y|,|Q_Y|)+μmin(|I_Y|,|Q_Y|),  Equation 4:whereEquation 5 Y=TAP_(BOC/QBOC) ^(w) ^(X) , simplifies the amplitudecalculation by eliminating the non-linear processing in Equation 3, andμ is a selected scale or constant scaling factor (e.g., 0.5). The firstdetector 201 and the second detector may use Equation 3 or Equation 4 todetermine or estimate the amplitude of the BOC or QBOC signalcomponents.

The decision-directed selection (DDSel) technique, as shown by Equation6, selects one component (BOC or QBOC) with more energy or signalamplitude for any given sampling period or interval, consistent with theDDSel amplitude plot of FIG. 9 that represents a universal amplitudefunction for any sampling period, where the universal correlationfunction includes the contributions from either the BOC or QBOCcorrelation functions that are associated with the greater signalamplitude during any sampling period or interval (e.g., an epoch).Accordingly, the DDSel code error 231, the DDSel carrier frequency error235 (fDDSel), and the DDSel carrier phase error 236 (φDDSel) result fromthe set of correlations (either BOC or QBOC) that has the greaterestimated amplitude for the sampling period. The amplitude and the errorcalculation results from the same set of correlations, regardless ofwhether it is based on BOC or QBOC; such selection of BOC or QBOCcorrelations guarantees appropriate normalization. Of theaccumulations/correlation data 125, the first detector 201 uses the Iand Q component for BOC correlation at prompt phase to estimate theprimary amplitude; the second detector 211 uses the I and Q componentfor QBOC correlation at prompt phase to estimate the secondaryamplitude.

In an alternate embodiment, where a multi-window BOC method is usedduring the pull-in instead of the DDSel method, the correlations for themulti-window-BOC (MWin-BOC) technique do not exceed the maximumcomponent selected by decision unit 35 or the data processor, whichguarantee the appropriate normalization. In one configuration theMWin-BOC refers to a dual EML error function. The horizontal axis ofFIG. 9 shows a time CD phase difference between the received signal andlocally generated replica signal in chips, whereas the vertical axisshows aggregate amplitude.

In one configuration, the tracking module 200 contains one or morediscriminators (202, 212, 203). The tracking module 200 comprises thecode tracking module 37 and the carrier tracking module 38. Adiscriminator (202, 212, 203) comprises a circuit, a set of correlators,or software instructions that can be adjusted to output, accept orreject signals of different characteristics, such as phase or frequency.In one embodiment, one or more discriminators (202, 212, 203), using thedifference between the early correlations and late correlations (or theEML correlation), synchronizes the received code with the locallygenerated PN code phase. The concentration of signal energy at the BOCcomponent indicates that CD tracking loop is closely tracking and lockedon the received signal, as opposed to distributing received signalenergy more evenly between the BOC and QBOC signal processing paths ofthe receiver. One or more discriminators (202, 212, 203) may facilitatedetermining a time offset or phase offset for the local generatedreplica with respect to the received CD phase to avoid trackingmultipath signals, phase noise, false lock, or a local peak correlationin the correlation function.

In FIG. 3, collectively FIG. 3A and FIG. 3B, the correlation data 125,including the I and Q vector of the E BOC, L BOC, E QBOC and L QBOCsignals, feed into a group of non-coherent, early-minus-late (EML)discriminators (202, 212, 203), where E and L represent early and late,respectively. As illustrated in FIG. 3, the group of non-coherentearly-minus-late (EML) discriminators (202, 212, 203) comprises: (a)first EML discriminator 202 (e.g., a BOC EML discriminator for a firstwindow (Window A or Wa)), (b) a second EML discriminator 212 (e.g., QBOCEML discriminator for a first window (Window A)), and (c) a third EMLdiscriminator 203 (e.g., a BOC EML discriminator for a second window(Window B or Wb)).

The first EML discriminator 202 and the second EML discriminator 212communicate with a first selection multiplexer 237. The first EMLdiscriminator 202 provides BOC-derived CD error 221 (e.g., code BOC Wa)to the first selection multiplexer 237. The second EML discriminator 212provides QBOC-derived CD error 222 to the first selection multiplexer237. Outputs of the first EML discriminator 202 and the third EMLdiscriminator 203 are summed by summer 275. The output (MWin-BOC signal)of the summer 275 and the output signal 231 (called cdDDSEL) areinputted into a multiplexer 243 that determines the code error signal233 based on the selection of DDSel method or MWin-BOC method for thepull-in mode, for example.

In an alternate embodiment of the pull-in mode, the multiplexer isconfigured to support a hybrid pull-in mode in which the DDSel method isused for tracking carrier phase and carrier frequency, in which theMWin-BOC method is used for code tracking, and in which the DDSelamplitude estimation is used for the DDSel tracking of carrier phase andcarrier frequency.

Depending on the selection signal 234 (SEL_MWin-BOC), the receivereither uses MWin-BOC method or DDSel method to pull in the CD phase. TheCD error signal 233 results from CD error 231 at DDSel mode; otherwise,the CD error signal 233 results from 232 at selection of MWin-BOC. Atevery sampling epoch, the CD error 231 uses either BOC-derived CD error221 or QBOC-derived CD error 222 depending on the selection signal(e.g., SEL_BOC) 123. MWin-BOC 232 combines two BOC-derived CD errors 221and 223, where CD error 221 and 223 result from BOC correlations withdifferent chip spacing. The output of the CD error 233, passing througha CD loop filter 263, generates a code rate signal 260 (e.g., codefrequency or code phase signal) for input to the CD NCO module 33.

As illustrated in FIG. 3 for the carrier tracking module 38, a group ofdelay modules (216, 217, 218, and 219) receives accumulations of theprompt BOC and QBOC signals for the in-phase (I) and quadrature phase(Q) vectors. The outputs of the delay modules provide input to a firstfrequency discriminator 204 and a second frequency discriminator 214.For example, the first frequency discriminator 204 uses the set of BOCcorrelations, whereas the second frequency discriminator 214 uses theset of QBOC correlations. The BOC-derived frequency error signal 225 andthe QBOC-derived frequency error signal 226 (e.g., fQBOC) are inputtedinto a second selection multiplexor 238, which outputs adecision-directed-selection frequency error signal 235 (e.g., fDDSel).The decision-directed (DDSel) method selects the BOC frequency error ifthe primary amplitude exceeds the secondary amplitude, otherwise itselects the QBOC frequency error to drive the CR loop.

Meanwhile, a first phase discriminator 205 and a second phasediscriminator 215 receive accumulations of the in-phase and quadraturevectors for the prompt BOC or QBOC signals. For example, the first phasediscriminator 205 uses the set of BOC correlations, whereas the secondphase discriminator 215 uses the set of QBOC correlations. The firstphase discriminator 205 outputs the CR phase error 227 (e.g., φBOC)derived from BOC correlations and the second phase discriminator 215outputs a carrier phase error 228 (e.g., φQBOC) derived from QBOCcorrelations. Similarly to frequency error selection, BOC-derived phaseerror 227 is selected if the primary amplitude is greater than thesecondary amplitude; otherwise, the QBOC-derived phase error 228 isselected to drive the CR loop via the decision-directed-selectioncarrier phase error 236. The carrier loop filter receives thedecision-directed-selection carrier phase error 236. The carrier ratesignal 270 from carrier loop filter 265 is applied to the carrier NCOmodule 34.

FIG. 4 illustrates the first detector 201 and the second detector 211 ofFIG. 3 in greater detail than FIG. 3. Like reference numbers in FIG. 3and FIG. 4 indicate like elements, blocks, processes, or sub-networks.

The first detector 201 comprises first set of absolute value modules350, a first minimum/maximum evaluator 320, a first scaler 377 (e.g.,first divider or a first multiplier by reciprocal value of the divisor(e.g., approximately 2)), and a first summer 353. The second detector211 comprises a second set of absolute value modules 351, a secondminimum/maximum evaluator 330, a second scaler 379 (e.g., second divideror multiplier by reciprocal value of the divisor (e.g., approximately2)), and a second summer 355. The first detector 201 and the seconddetector 211 may use Equation 4 to compute the BOC and QBOC amplitudes.

In the first detector 201, the first set of absolute value modules 350determines the absolute value of the prompt IBOC signal 301 and theprompt QBOC signal 311. The absolute values (302, 312) of the promptIBOC signal and the prompt QBOC signal are inputted into theminimum/maximum evaluator 320 that determines the maximum aggregatesignal amplitude 303 of the prompt IBOC signal and the prompt QBOCsignal and the minimum aggregate signal amplitude 313 of the prompt IBOCsignal and the prompt QBOC signal. By application of the scaler 377, theminimum aggregate signal amplitude 313 is scaled approximately byone-half at every sampling period. The output of the scaler 377 is addedto the maximum aggregate amplitude 303 in the summer 353, which outputsthe prompt BOC amplitude (ABOC) 305 for input into the decision unit 35.In one embodiment, the scaler 377 may comprise a multiplier, a divider,an amplifier or another suitable device.

In the second detector 211, the second set of absolute value modules 351determines the absolute value of the prompt IQBOC signal 321 and theprompt QQBOC signal 331. The absolute values (322, 332) of the promptIQBOC signal and the prompt QQBOC signal are inputted into theminimum/maximum evaluator 330 that determines the maximum aggregatesignal amplitude 323 of the prompt IQBOC signal and the prompt QQBOCsignal and the minimum aggregate signal amplitude 333 of the promptIQBOC signal and the prompt QQBOC signal. By application of the scaler379, the minimum aggregate signal amplitude 333 is scaled approximatelyby half at every sampling period. The output of the scaler 379 is addedto the maximum aggregate amplitude 323 in the summer 355, which outputsthe prompt QBOC amplitude (AQBOC) 325 for input to the decision unit 35.In one embodiment, the scaler 379 may comprise a multiplier, a divider,an amplifier or another suitable device.

In one embodiment, the decision unit 35 or data processor may use thefollowing equation:A _(DDSel)=max(Amp(P _(BOC) ^((0,1))),Amp(P _(QBOC) ^((0,1)))  Equation6:

In one embodiment, the decision unit 35 or data processor calculates theoverall signal amplitude 240 (e.g., A_(DDSel)). Equation 6 may includean approximation of Equation 4 as a substitute for the ideal envelopecalculation of Equation 3. The first detector 201 and the seconddetector 211 may use Equation 3 or Equation 4 to determine or estimatethe amplitude of the BOC or QBOC signal components. Further Equation 6provides a selection signal 210 to the logic device 271 (e.g., OR gateor exclusive-OR gate) through comparing the primary amplitude with thesecondary amplitude. The output of logic device 271 provides the BOCselection signal 123 (e.g., SEL_BOC).

As mentioned before, to achieve reliable signal detection, smallersearch spacing is required to avoid multiple zero-crossing points on theBOC correlation. Such demand extends the acquisition time, and requireseither software, hardware, or both to finely control the phase of localreplica. As illustrated by the universal correlation function of FIG. 9,the DDSel method, eliminates the zero-crossing point within −1 and +1chips of the peak amplitude and associated prompt correlation. Theuniversal correlation function of FIG. 9 supports computationalstability and correct estimation of signal power.

FIG. 5A and FIG. 5B are collectively referred to as FIG. 5. FIG. 5 showsthe first EML discriminator 202, the second EML discriminator 212, andthe third EML discriminator 203 in more detail than FIG. 3. Likereference numbers in FIG. 3 and FIG. 5 indicate like elements.

The first EML discriminator 202 receives input correlation data,accumulation data or other input data (401, 402, 411, 412). The firstEML discriminator 202 comprises an early (E) BOC envelope/amplitudedetector 405 and a late (L) BOC envelope/amplitude detector 415 thatprovide outputs (E BOC amplitude 406 and L BOC amplitude 416) toconstruct a first EML accumulation at summer output 409 through summer451. At the summer output of summer 451, the first EML accumulation isthen normalized by normalizer 490 (e.g., normalization unit) to providea BOC-derived CD error 221 with first window of Wa to first selectionmultiplexer 237. In one configuration, the early BOC envelope detector405 and the late BOC envelope detector 415 may comprise the same orsimilar software modules or the same or similar algorithms that areapplied to early BOC correlation data and late BOC correlation data,respectively.

The second EML discriminator 212 receives input correlation data,accumulation data or other input data (421, 422, 431, 432). The secondEML discriminator 212 comprises an early (E) QBOC envelope/amplitudedetector 425 and a late (L) QBOC envelope/amplitude detector 435 thatprovide outputs (E QBOC amplitude 426 and L QBOC amplitude 436) toconstruct a second EML accumulation at summer output 429 through summer453. The second EML accumulation is then normalized by normalizer 491(e.g., normalization unit) to provide a code QBOC correlation as asecond QBOC-derived CD error 222 with first window of Wa to firstselection multiplexer 237. In one configuration, the early QBOC envelopedetector 425 and the late QBOC envelope detector 435 may comprise thesame or similar software modules or the same or similar algorithms thatare applied to early QBOC correlation data and late QBOC correlationdata, respectively.

The third EML discriminator 203 receives input correlation data,accumulation data or other input data (451, 452, 461, 462). The thirdEML discriminator 203 comprises an early (E) BOC envelope/amplitudedetector 455 and a late (L) BOC envelope/amplitude detector 465 thatprovide outputs (E BOC amplitude 456 with Wb and L BOC amplitude 466with Wb) at nodes to construct a third EML accumulation at summer output459 through summer 457. The third EML accumulation is then normalized bynormalizer 492 to provide another BOC-derived code error 223 (e.g.,cdBOC Wb) with second window of Wb, where the first window (Wa) does notequal the second window (Wb). In one configuration, the early BOCenvelope detector 455 and the late QBOC envelope detector 465 maycomprise the same or similar software modules or the same or similaralgorithms that are applied to early QBOC correlation data and late QBOCcorrelation data, respectively.

In FIG. 5, the decision-directed selection, early-minus-late code error(DDSel EML CD) error results from either first EML discriminator 202 orsecond EML discriminator 212 depending on the BOC selection signal 123.Such DDSel method removes the false lock points on the discriminatorS-curve. The discriminator S-curve means an early minus late correlationfunction produced by the difference between an early correlator and alate correlator with respect to the received signal. If BOC selectionsignal 123 is a first logic level (e.g., high logic level or logic level1), the first EML discriminator 202 enables the amplitude calculation ofP_(BOC) ^((0,Wa)) using module 405 and the amplitude calculation ofP_(BOC) ^((1-Wa,1)) using the envelope detector 415 of FIG. 5. Thedifference between P_(BOC) ^((0,Wa)) and P_(BOC) ^((1-Wa,1)) are thennormalized by A_(DDSel) to generate the BOC-derived, DDSel EML CD error221 at the output of the first EML discriminator 202.

If the BOC selection signal 123 is a second logic level (e.g., low logiclevel or 0 logic level), which is different than the first logic level,the second EML discriminator 212 enables the amplitude calculation ofP_(QBOC) ^((0,Wa)) using envelope detector 425 and the amplitudecalculation of P_(QBOC) ^((1-Wa,1)) using envelope detector 435. Thedifference between P_(QBOC) ^((0,Wa)) and P_(QBOC) ^((1-Wa,1)) are thennormalized by A_(DDsel) to generate the QBOC-derived, DDSel EML CD error222 (e.g., cdQBOC Wa) at the output of the second EML discriminator 212.The normalized code error is modeled by Equation 7.

$\begin{matrix}{{\tau = \frac{{E_{xBOC}} - {L_{xBOC}}}{A_{DDSel}}},} & {{Equation}\mspace{14mu} 7}\end{matrix}$referring to FIG. 5, where

-   -   xBOC is either BOC or QBOC,    -   E_(xBOC) corresponds to P_(xBOC) ^((0,Wa)),    -   L_(xBOC) corresponds to P_(xBOC) ^((1-Wa,1)),        where the DDSel reveals computation advantage in software        implementation as only one component (either BOC or QBOC) really        needs to be calculated.

Alternatively to the DDsel mode of operation during pull-in of the codeof the received composite signal, a multi-window BOC code mode may beused during pull-in of the code. For example, a linear combination usingthe first BOC-derived code error 221 based on the first window (Wa) andthe third BOC-derived code error 223 based on the second window (Wb) caneliminate the false zero-crossing points on discriminator S-curve, wherethe chip spacing for error signal 221 is different from the chip spacingfor error signal 223, consistent with the first window and the secondwindow. As shown in FIG. 5, a block diagram of the linear combinationincludes two generally linear scaling devices 471 and 472, and acombiner 475 (e.g., summer). The resultant error estimation 232 (e.g.,composite code error estimate) is named MWin-BOC code error modeledthrough Equation 8.

$\begin{matrix}{{\tau_{{MWin}\text{-}{BOC}} = {{\alpha\frac{{P_{BOC}^{({0,{Wa}})}} - {P_{BOC}^{({{1 - {Wa}},1})}}}{A_{DDSel}}} + {\beta\;\frac{{P_{BOC}^{({0,{W\; b}})}} - {P_{BOC}^{({{1 - {W\; b}},1})}}}{A_{DDSel}}}}},} & {{Equation}\mspace{14mu} 8}\end{matrix}$referring to FIG. 5, where

-   -   α is the linear gain (e.g., at mixer or multiplier 472) applied        on the first BOC-derived code error estimation with window Wa,    -   β is the linear gain (e.g., at mixer or multiplier 471) applied        on the secondary BOC-derived code error estimation with window        Wb.

FIG. 6 discloses the nonlinear estimation on the CR frequency and CRphase estimation. Like reference numbers in FIG. 1, FIG. 3 and FIG. 6indicate like elements. The tracking module 200 or the carrier trackingmodule 38 or the carrier phase/frequency error module 45 may conduct thenonlinear estimation on the CR frequency and CR phase estimation. Theblock diagram of FIG. 6 shows a first frequency discriminator 204, asecond frequency discriminator 214, a first phase discriminator 205, anda second phase discriminator 215.

As shown in FIG. 6, the first frequency discriminator 204 receives inputdata, such as prompt IBOC signal 501 and prompt QBOC signal 511. Similarto the CR phase estimation previously mentioned in this document, thefirst frequency discriminator 204, for BOC-derived CR frequency errordetection, comprises two delay units (502, 512). The two delay units(502, 512) store the BOC correlations (503, 513) for the previoussampling period. Two multipliers (504, 514) remove the CD uncertaintyinvolved in the BOC correlations. A summer 508 to form a delta CR phaseobservation 506 over a sampling period, and a normalizer 507 (e.g.,normalization unit) to remove the amplitude information modulated on thesignal 506. The normalizer 507 outputs CR frequency error 225 (e.g.,fBOC), derived from BOC correlations.

The second frequency discriminator 214 receives input data, such asprompt IQBOC signal 521 and prompt QQBOC signal 531. Like BOC-derivedfrequency error, a QBOC CR frequency error 226 is generated using thesecond frequency discriminator 214 in FIG. 6. The second frequencydiscriminator 214, for QBOC-derived CR frequency error detection,comprises two delay units (522, 532). The two delay units (524, 534)store the QBOC correlations (523, 533) for the previous sampling period.Two multipliers (524, 534) remove the CD uncertainty involved in theQBOC correlations. A summer 528 forms a delta CR phase observation 526over a sampling period, and a normalizer 527 (e.g., normalization unit)removes the amplitude information modulated on the signal 526. Thenormalizer 527 outputs CR frequency error frequency error 226 (e.g.,fQBOC), derived from QBOC correlations.

Depending on the BOC selection signal 123, the digital receiver portion192 or the tracking module 200 uses either the CR frequency error 225,(e.g., fBOC) derived from BOC correlations, or the CR frequency errorfrequency error 226 (e.g., fQBOC), derived from QBOC correlations, todrive the CR loop.

For a general BOC correlation, modeled by Equation 1, both I componentand Q component are convolved with CD uncertainty of 2πN_(BOC)τ and CRphase uncertainty of δφ. In order to extract the CR phase error δφ fromthe set of BOC correlations, the CD uncertainty of 2πN_(BOC)τ needs tobe wiped off. Equation 9 shows that the cross-product CR phase detection544, using multiplier 542, can remove the code uncertainty.

$\begin{matrix}{{{QP}_{BOC}{IP}_{BOC}} = {A^{2}\left\{ {R^{2}\left( \tau_{k} \right)} \right\}\left\{ {\cos^{2}\left( {2\pi\; N_{BOC}\tau_{k}} \right)} \right\}\left\{ {\frac{1}{2}\sin\; 2\varphi_{k}} \right\}}} & {{Equation}\mspace{14mu} 9}\end{matrix}$

Similarly, for a general QBOC correlation, modeled by Equation 2, both Icomponent and Q component are convolved with CD uncertainty of2πN_(BOC)τ and CR phase uncertainty of δφ. Equation 10 shows that thecross-product CR phase detection 554, using multiplier 552, can removethe CD uncertainty.

$\begin{matrix}{{{QP}_{QBOC}{IP}_{QBOC}} = {A^{2}\left\{ {R^{2}\left( \tau_{k} \right)} \right\}\left\{ {\sin^{2}\left( {2\pi\; N_{BOC}\tau_{k}} \right)} \right\}\left\{ {\frac{1}{2}\sin\; 2\varphi_{k}} \right\}}} & {{Equation}\mspace{14mu} 10}\end{matrix}$

In FIG. 6, the BOC-derived CR phase error 227 is obtained by thenormalizer 547 normalizing the BOC-derived phase detection 544 by squareof amplitude using DDSel method. Similarly, the QBOC-derived CR phaseerror 228 is achieved by the normalizer 557 normalizing the QBOC-derivedphase detection 554 by square of amplitude using DDSel method. The BOCselection signal 123 determines whether BOC-derived CR phase error orthe QBOC-derived CR phase error is used to drive the CR loop. The BOCselection signal 123 controls the multiplexers (238, 239).

FIG. 7A generally discloses a method for receiving a composite signal inaccordance with any embodiments of the receiver (e.g., 11) disclosed inFIG. 1 through FIG. 6, inclusive. In particular, FIG. 7A discloses amethod, operating at pull-in mode, to acquire the code and carrier for areceived composite signal. FIG. 7A illustrate the use of decision-directselection (DDSel) to drive the code tracking, carrier tracking, andamplitude estimation. The method of FIG. 7A begins in block S700.

In step S700, a receiver 11 (e.g., satellite navigation receiver) or adigital receiver portion 192 receives a binary offset carrier (BOC)modulated signals to extract a BOC component by mixing or combining witha local BOC replica, and to derive a quadrature BOC (QBOC) component bycombining with a local QBOC replica. In one embodiment, the BOCcomponent comprises an in-phase BOC component and a quadrature-phase BOCcomponent and the QBOC component comprises in-phase QBOC component and aquadrature-phase QBOC component. For example, the receiver 11 comprisesa code mixer 42 (FIG. 1) for mixing or combining the BOC signal with thelocal BOC or QBOC replica from one or more signal generators (e.g.,multiplexed output of a first signal generator 32 and second signalgenerator 31).

In step S702, a first detector 201 (e.g., in FIG. 1, FIG. 3 and FIG. 4)or data processor detects a primary amplitude of the BOC componentduring a sampling period or interval. Step S702 may be executed inaccordance with various techniques, which may be applied alternately orcumulatively.

Under a first technique, the first detector 201 or data processordetects or measures the signal energy or aggregate power of the BOCcomponent comprising in-phase BOC component and a quadrature-phase BOCcomponent.

Under a second technique, the first detector 201 or data processorapplies the correlation P_(BOC) ^((0,1)) of Equation 3 to provide anamplitude (e.g., an ideal amplitude) estimate for a BOC component of thereceived composite signal during a sampling period or a time interval.For example, the first detector 201 applies a correlation P_(BOC)^((0,1)) of Equation 3 to provide ideal amplitude estimation for BOCcomponent: Amp(Y)=√{square root over (I_Y²+Q_Y²)},

where

I_Y is the in-phase BOC component, and

Q_Y is the quadrature-phase BOC component.

Under a third technique, the first detector 201 or the data processeruses the linear approximation Equation 4 to simplify or replaces thecalculation of Equation 3. In one embodiment, the linear approximationof Equation 4 has a bias that is acceptable and facilitates rapidestimation of the primary amplitude. Under the third technique, theestimating or detecting of the primary amplitude or the secondaryamplitude (e.g., by detectors 201, 211) for the sampling period is basedon the following equations that use linear approximation separately forthe BOC and QBOC signal components:Amp^(approx)(Y)

max(|I_Y|,|Q_Y|)+μmin(|I_Y|,|Q_Y|),where

Y=P_(BOC) ^(W) ^(X) ,

μ is a selected scale or a constant scaling factor (e.g., 0.5),

W_(x) is the window size of the correlator,

I_Y is the in-phase xBOC component, and

Q_Y is the quadrature-phase xBOC component,

xBOC refers to the BOC or QBOC signal component of the receivedcomposite signal. In the above equation, for instance, the firstdetector 201 detects the BOC signal component and substitutes the BOCsignal component for xBOC.

In step S704, a second detector 211 (in FIG. 1, FIG. 3 and FIG. 4) orthe data processor detects a secondary amplitude of the QBOC componentduring a sampling period or interval. Step S704 may be executed inaccordance with various techniques, which may be applied alternately orcumulatively.

Under a first technique, the second detector 211 or data processordetects or measures the signal energy or aggregate power of the QBOCcomponent comprising in-phase QBOC component and a quadrature-phase QBOCcomponent.

Under a second technique, the second detector 211 or data processorapplies a correlation P_(QBOC) ^((0,1)) of Equation 3 to provide idealamplitude estimation for QBOC component: Amp(Y)=√{square root over(I_Y²+Q_Y²)}, where

I_Y is the in-phase QBOC component, and

Q_Y is the quadrature-phase QBOC component.

Under a third technique, the linear approximation Equation 4 simplifiesor replaces the calculation of FIG. 3. In one embodiment, the linearapproximation of Equation 4 has a bias that is acceptable andfacilitates rapid estimation of the primary amplitude. Under the thirdtechnique, the estimating or detecting the secondary amplitude (e.g., bythe second detector 211 or data processor) for the sampling period isbased on the following equations that use linear approximationseparately for the QBOC signal components:Amp^(approx)(Y)

max(|I_Y|,|Q_Y|)+μmin(|I_Y|,|Q_Y|),where

Y=P_(QBOC) ^(W) ^(X) ,

μ is a selected scale or a constant scaling factor (e.g., 0.5),

W_(x) is the window size of the correlator,

I_Y is the in-phase QBOC component, and

Q_Y is the quadrature-phase QBOC component.

In step S706, a decision unit 35, electronic data processor, or digitalreceiver portion 192 determines whether or not the primary amplitudeexceeds (or equals) the secondary amplitude for a sampling period. Forexample, in step S706 does the primary amplitude exceed the secondaryamplitude, or is the primary amplitude greater than or equal to thesecondary amplitude? In step S706, if the primary amplitude exceeds orequals the secondary amplitude, the method continues with step S708.However, if the primary amplitude does not exceed the secondaryamplitude, the method continues with step S710.

In step S708, data processor, a selector, or one or more multiplexers(e.g., second multiplexer 29 for each received channel) select a set ofBOC correlations (e.g., early, prompt and late BOC correlations) inaccordance with a BOC correlation function for the sampling period ifthe primary amplitude exceeds or equals the secondary amplitude for thesampling period. For example, the second multiplexer 29 selects a set ofBOC correlations (e.g., via the BOC selection signal 123 provided tomultiplexer 29) in accordance with a BOC correlation function for thesampling period if the primary amplitude exceeds or equals the secondaryamplitude for the sampling period. In one embodiment, the zero crossingpoint for the BOC correlation function offsets by approximately

$\frac{1}{4N_{BOC}}$against the zero crossing point for the QBOC correlation function.

In one possible configuration in step S708, data processor, a selector,or one or more multiplexers (e.g., second multiplexer 29 for eachreceived channel) select a first correlation or first set of BOCcorrelations associated with or resulting from: (a) substantially prompttiming (e.g., prompt signal 117) of the first signal generator 32 withrespect to the received digital signal (102) and (b) a substantiallyfull chip window (e.g., established by the code phase window module 157)within the correlator module 130.

In one possible configuration for step S708, the set of BOC correlationsis modeled or estimated by the following equation:

$\begin{matrix}{{TAP}_{BOC}^{w_{X}} = {{AW}_{X}{\sum\limits_{{i = 0},{{di} = {F_{Chip}/F_{S}}}}^{N_{Chip}}\left\{ \begin{matrix}\begin{matrix}{c_{i}{c_{i - \tau}.}} \\{{\sin\left( {2\pi\; N_{BOC}i} \right)}{{\cos\left( {2\pi\;{N_{BOC}\left( {i - \tau} \right)}} \right)}.}}\end{matrix} \\{\mathbb{e}}^{j\;\delta\;\theta_{i}}\end{matrix} \right.}}} \\{= {\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}{\mathbb{e}}^{j\delta\varphi}}} \\{{= {\overset{\overset{{I\_ TAP}_{BOC}^{W_{X}}}{︷}}{\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}\cos\;{\delta\varphi}} + {j\underset{\underset{{Q\_ TAP}_{BOC}^{W_{X}}}{⎴}}{\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}\sin\;\delta\;\varphi}}}},}\end{matrix}$where

-   -   TAP_(BOC) ^(w) ^(X) is the correlation based on the BOC input        and BOC local replica with a selection of PN TAP, where PN TAP        can be early, prompt or late,    -   i is chip phase in unit of chips,    -   N_(Chip) is number of chips per millisecond,    -   di is sample phase increment in unit of chips,    -   A is the amplitude of signal at level of millisecond        integration,    -   W_(X) is the window size in chips,    -   N_(BOC) is equal to m/n of BOC (m,n), where m is f_(m)/f_(c) and        n is f_(n)/f_(c), f_(m) is a first subcarrier frequency, f_(n)        is the actual chip frequency, and f_(c) is the reference        chipping rate,    -   F_(Chip) is the chipping rate in chips/second,    -   F_(S) is the sampling rate in samples/second,    -   R(τ) is the conventional correlation function for PN code,    -   τ is the chip phase error estimated at receiver side,    -   δθ_(i) is the carrier phase estimation error for each sample,    -   δφ is the average carrier phase estimation error over N_(Chip)        period,    -   I_TAP_(BOC) ^(w) ^(X) is the in-phase or real part of TAP_(BOC)        ^(w) ^(X) , and    -   Q_TAP_(BOC) ^(w) ^(X) is the quadrature or imaginary part of        TAP_(BOC) ^(w) ^(X) .

The digital receiver portion 192 or code tracking module 37 can useabove equation for determining the correlation of step S708 to formearly-minus-late functions for estimating the code error for thereceived composite signal in step S711.

In step S710, if the primary amplitude is less than the secondaryamplitude or if the secondary amplitude exceeds the primary amplitude,the data processor, selector, or one or more multiplexors (e.g., secondmultiplexer 29 for each received channel) select a set of QBOCcorrelations (e.g., early, prompt and late QBOC correlations) inaccordance with a QBOC correlation function for current sampling periodto generate the error estimations to drive the code (CD) and carrier(CR) feedback loop. In one embodiment, the zero crossing point for theQBOC correlation function offsets by approximately

$\frac{1}{4N_{BOC}}$against the zero crossing point for the BOC correlation function.

In one possible configuration in step S710, data processor, a selector,or one or more multiplexers (e.g., second multiplexer 29 for eachreceived channel) select a second correlation or second set of QBOCcorrelations associated with or resulting from: (a) substantially prompttiming (e.g., prompt signal 117) of the first signal generator 32 withrespect to the received digital signal (102) and (b) a substantiallyfull chip window (e.g., established by the code phase window module 157)within the correlator module 130.

Under one possible configuration in step S710, a set of QBOCcorrelations is modeled by the following equation:

$\begin{matrix}{{TAP}_{QBOC}^{w_{X}} = {{AW}_{X}{\sum\limits_{{i = 0},{{di} = {F_{Chip}/F_{S}}}}^{N_{Chip}}\left\{ \begin{matrix}\begin{matrix}{c_{i}{c_{i - \tau}.}} \\{{\sin\left( {2\pi\; N_{BOC}i} \right)}{{\cos\left( {2\pi\;{N_{BOC}\left( {i - \tau} \right)}} \right)}.}}\end{matrix} \\{\mathbb{e}}^{{j\theta}_{i}}\end{matrix} \right.}}} \\{= {\frac{A}{2}W_{X}{R(\tau)}{\sin\left( {2\pi\; N_{BOC}\tau} \right)}{\mathbb{e}}^{j\delta\varphi}}} \\{{= {\overset{\overset{{I\_ TAP}_{QBOC}^{W_{X}}}{︷}}{\frac{A}{2}W_{X}{R(\tau)}{\sin\left( {2\pi\; N_{BOC}\tau} \right)}\cos\;{\delta\varphi}} + {j\underset{\underset{{Q\_ TAP}_{QBOC}^{W_{X}}}{⎴}}{\frac{A}{2}W_{X}{R(\tau)}{\sin\left( {2\pi\; N_{BOC}\tau} \right)}\sin\;\delta\;\varphi}}}},}\end{matrix}$where

-   -   TAP_(BOC) ^(w) ^(X) is the correlation based on the BOC input        and QBOC local replica with a selection of PN TAP, where PN TAP        can be early, prompt or late,    -   i is chip phase in unit of chips,    -   N_(Chip) is number of chips per millisecond,    -   di is sample phase increment in unit of chips,    -   A is the amplitude of signal at level of millisecond        integration,    -   W_(X) is the window size in chips,    -   N_(BOC) is equal to m/n of BOC (m,n), where m is f_(m)/f_(c) and        n is f_(n)/f_(c), f_(m) is a first subcarrier frequency, f_(n)        is the actual chip frequency, and f_(c) is the reference        chipping rate,    -   F_(Chip) is the chipping rate in chips/second,    -   F_(S) is the sampling rate in samples/second,    -   R(τ) is the conventional correlation function for PN code,    -   τ is the chip phase error estimated at receiver side,    -   δθ_(i) is the carrier phase estimation error for each sample,    -   δφ is the average carrier phase estimation error over N_(Chip)        period,    -   I_TAP_(QBOC) ^(w) ^(X) is the in-phase or real part of        TAP_(QBOC) ^(w) ^(X) , and    -   Q_TAP_(QBOC) ^(w) ^(X) is the quadrature or imaginary part of        TAP_(QBOC) ^(w) ^(X) .

The digital receiver portion 192 or code tracking module 37 can useabove equation for determining the correlation of step S710 to formearly-minus-late functions for estimating the code error for thereceived composite signal in step S711.

In step S711, the data processor, receiver 11 or tracking module 200processes the selected correlations (e.g., from step S708 or step S710for each sampling period) to track code and carrier (e.g., carrierfrequency, carrier phase or both) of the received composite signal forthe estimation of a range between a receiver antenna and a satellitetransmitter that transmits the received composite signal. A measurementgeneration module 39 or a navigation positioning engine 41 can estimatethe range or position of the receiver antenna, for example.

Step S711 may be carried out in accordance with various techniques thatmay be applied separately or cumulatively. Under a first technique, thedata processor or digital receiver portion 192 uses the BOC-derivederror for tracking code error, frequency error and phase error for asampling period (e.g., epoch) if the primary amplitude exceeds, orequals, the secondary amplitude. Accordingly, the data processor orreceiver 11 uses the selected set of BOC correlations in accordance witha BOC correlation function for a current sampling period to generate theerror estimations to drive CD and CR feedback loop.

Under a second technique, the data processor or digital receiver portion192 uses the QBOC-derived error for tracking code error, frequency errorand phase error of a sampling period if the second amplitude exceeds theprimary amplitude. Accordingly, the data processor or receiver uses theselected set of QBOC correlations in accordance with a QBOC correlationfunction for a current sampling period to generate the error estimationsto drive CD and CR feedback loop.

Under a third technique, for each successive sampling period in a codeloop during a pull-in mode that used DDSel, the data processor, trackingmodule 200, or code tracking module 37 selects or processes the set (orsubset) of BOC correlations (e.g., early and late BOC correlations) orthe set (or subset) of QBOC correlations (e.g., early and late QBOCcorrelations) to form the early-minus-late estimation of the code errorτ based on greater amplitude resulting from the BOC component or fromthe QBOC component, respectively, which is modeled by the followingequation:

${\tau = \frac{{E_{xBOC}} - {L_{xBOC}}}{A_{DDSel}}},$where

-   -   xBOC is either BOC or QBOC,    -   E_(xBOC) corresponds to P_(xBOC) ^((0,Wa)),    -   L_(xBOC) corresponds to P_(xBOC) ^((1-Wa,1)), and        where W_(a) is window size of the correlator. For example, in        the code tracking module 37 the code multiplexer 237 is        controlled to output or select a BOC-derived CD error signal        (231) or QBOC-derived CD error signal (231) via the BOC        selection signal 123 provided from the decision unit 35, or        through the logic device 271.

Under a fourth technique, for each successive sampling period in acarrier phase loop, the xBOC correlations (e.g., prompt BOC or promptQBOC correlations), with greater amplitude, form the carrier phase errorestimation, wherein the cross-product using the in-phase and quadraturephase component resolves the ambiguity from sine or cosine of BOCmisalignment.

Under a fifth technique, the digital receiver portion 192 or dataprocessor processes selected prompt BOC correlations or selected promptQBOC correlations during the sampling period, or both correlations oversuccessive sampling periods, to track a carrier of the receivedcomposite signal, or track both the carrier and the frequency of thereceived composite signal.

FIG. 7B refers to FIG. 7B-1 and FIG. 7B-2 collectively. FIG. 7Bgenerally illustrates a flow chart of a method for receiving a compositesignal comprising a receiver for receiving a composite signal. Inparticular, FIG. 7B provides an illustration of BOC tracking strategy inboth pull-in mode and steady state mode. Like reference numbers in FIG.7A and FIG. 7B indicate like elements, steps or procedures.

The method of FIG. 7B begins in block S700. Steps 700, 702, and 704 weredescribed in FIG. 7A and the above description applies equally to FIG.7B as if fully set forth here.

In the illustrative example of FIG. 7B, step 716 follows step S704. Instep 716, the digital receiver portion 192, the data processor, or acombination of the decision module 35 and the decision augmentationmodule 220 (in FIG. 3A) determines whether (or not) the receiveroperates in a steady state mode. For the received composite BOC signal,if the primary amplitude continuously exceeds the secondary amplitudefor a threshold number of consecutive sampling periods or M epochs, thereceiver (11) or the digital receiver portion 192 switches to or isdetermined to be in the steady state mode, otherwise the receiverremains in or is determined to be in the pull-in mode. In oneconfiguration, the threshold number or M may be any whole number orinteger greater than (3) three. However, the threshold number or M maybe any suitable number that is determined by empirical evidence,operational tests, factory setting, programmable setting or otherwise ofthe receiver that reliably indicates operation of the receiver in thesteady state mode for one or more received channels of the composite BOCsignal.

The pull-in mode is a pre-alignment state in which the receiver isattempting to align the code, phase and frequency of the local replicasignal with the received composite signal to track and demodulateeffectively, reliably the received composite signal. In the pull-in modethe demodulated energy may be divided between the BOC component and QBOCcomponent of the received signal, making it more difficult toaccurately, reliably recover or decode the modulation on the receivedcomposite signal. In contrast, in the steady state mode, the demodulatedenergy is mostly concentrated in the BOC component.

If the digital receiver portion 192, the data processor, or thecombination of the decision unit 35 and the decision augmentation module220 determine that that the receiver (e.g., one or more channels of thereceived composite signal) is operating in the steady state mode, themethod continues with step S718. However, if the digital receiverportion 192, data processor or the combination of the decision unit 35and the decision augmentation module 220 determine that the receiver isnot operating in the steady state mode, the method continues with stepS706.

In step S718, the digital receiver portion 192, the code tracking module37 or the tracking module 200 processes BOC correlations (e.g., selectedBOC correlations or steady state BOC correlations collected duringsampling periods of the steady state mode) by determining a code errorin accordance with a suitable early-minus-late function for the steadystate mode (e.g., coherent discriminator function). Further, the digitalreceiver portion, the code tracking module 37 or the tracking module 200process the BOC correlations (e.g., selected BOC correlations or otherBOC correlations) by determining a carrier phase error in accordancewith a Costas or Phase Locked Loop (PLL) phase error function. The BOCcorrelations may be collected over a single window of the correlatormodule 130, for example.

In one possible configuration for carrying out step S718, the digitalreceiver portion 192, the code tracking module 37 or the tracking module200 selects or processes only the BOC correlations for the steady statemode and uses only a BOC error function during the steady state mode foran epoch during the steady state mode, unless a cycle slip occurs. Acycle slip occurs when there is a discontinuity in measured carrierphase or the number of measured wavelengths between a satellite and thereceiver from a temporary loss of lock of the carrier tracking loopwithin the carrier loop module 46 or the tracking module 200.

In the steady state mode, the demodulated energy is mostly concentratedin the BOC component. Therefore, in the steady state mode the receiver,data processor, or the decision unit 35 can disregard the QBOC componentof the received composite signal under such circumstances, mitigate thenoise impact and thus improve the tracking accuracy of the trackingmodule 200. As shown in FIG. 7B, step S718 only uses the BOCcorrelations, identical to tracking any binary phase shift keying (BPSK)signal, to drive a code tracking module 37, and to drive the carriertracking module 38.

In step S706, a decision unit 35, electronic data processor, or digitalreceiver portion 192 determines whether or not the primary amplitudeexceeds (or equals) the secondary amplitude for a sampling period. Forexample, in step S706 does the primary amplitude exceed the secondaryamplitude, or is the primary amplitude greater than or equal to thesecondary amplitude? In step S706, if the primary amplitude exceeds orequals the secondary amplitude, the method continues with step S708.However, if the primary amplitude does not exceed the secondaryamplitude, the method continues with step S710.

In step S708, data processor, a selector, or one or more multiplexers(e.g., second multiplexer 29 for each received channel) select a set ofBOC correlations in accordance with a BOC correlation function for thesampling period if the primary amplitude exceeds or equals the secondaryamplitude for the sampling period. For example, the second multiplexer29 selects a set of BOC correlations (e.g., via the BOC selection signal123 provided to multiplexer 29) in accordance with a BOC correlationfunction for the sampling period if the primary amplitude exceeds orequals the secondary amplitude for the sampling period.

In one possible configuration in step S708, data processor, a selector,or one or more multiplexers (e.g., second multiplexer 29 for eachreceived channel) select a first correlation or first set of BOCcorrelations associated with or resulting from: (a) substantially prompttiming (e.g., prompt signal 117) of the first signal generator 32 withrespect to the received digital signal (102) and (b) a substantiallyfull chip window (e.g., established by the code phase window module 157)within the correlator module 130.

In step S710, if the primary amplitude is less than the secondaryamplitude or if the secondary amplitude exceeds the primary amplitude,the data processor, selector, or one or more multiplexors (e.g., secondmultiplexer 29 for each received channel) select a set of QBOCcorrelations in accordance with a QBOC correlation function for currentsampling period to generate the error estimations to drive the code (CD)and carrier (CR) feedback loop. In one embodiment, the zero crossingpoint for the QBOC correlation function is offset by approximately

$\frac{1}{4N_{BOC}}$against the zero crossing point for the BOC correlation function.

In one possible configuration in step S710, data processor, a selector,or one or more multiplexers (e.g., second multiplexer 29 for eachreceived channel) select a second correlation or second set of QBOCcorrelations associated with or resulting from: (a) substantially prompttiming (e.g., prompt signal 117) of the first signal generator 32 withrespect to the received digital signal (102) and (b) a substantiallyfull chip window (e.g., established by the code phase window module 157)within the correlator module 130.

In the pull-in mode, step S712 follows step S708. In step S712, thedigital receiver portion 192, the data tracking module 200, the codetracking module 37, or the carrier tracking module 38 process theselected BOC correlations. Step S712 may be carried out by executing oneor more procedures that may be applied separately or cumulatively.

Under a first procedure for executing step S712, the data processor ordata tracking module 200 determines a first code error in accordancewith a first early-minus-late function.

Under a second procedure, the data processor or carrier tracking module200 determines a first carrier frequency error in accordance with afrequency error function (e.g., first frequency error function).

Under a third procedure, the data processor or carrier tracking module200 determines a first carrier frequency error in accordance with adot-product, frequency error function (e.g., first frequency errorfunction) using a prompt BOC correlation and a prompt BOC correlation ofa previous epoch or previous sampling period.

Under a fourth procedure, the data processor or carrier tracking module38 determines a first carrier phase error in accordance with a phaseerror function (e.g., first phase error function).

Under a fifth procedure, the data processor or carrier tracking module38 determines a first carrier phase error in accordance with adot-product, phase error function (e.g., first phase error function)using the prompt BOC correlation.

Under a sixth procedure, any of the above procedures of step S712 areexecuted if the primary amplitude exceeds the secondary amplitude forthe sampling period or if the primary amplitude equals the secondaryamplitude for the sampling period. Under the seventh procedure, thefirst early-minus-late function comprises a BOC early-minus-latefunction that has a chip spacing of approximately 0.4 chips.

Step S714 follows step S710. In step S714, the digital receiver portion192, the tracking module 200, the code tracking module 37, or thecarrier tracking module 38 process the selected QBOC correlations. StepS714 may be carried out by executing one or more procedures that may beapplied separately or cumulatively.

Under a first procedure for executing step S714, the data processor ordata tracking module 200 determines a second code error in accordancewith a second early-minus-late function or another early-minus-lateerror function. In certain embodiments, the second early-minus-latefunction is distinct from the first early-minus-late function (e.g., ofstep S712), whereas in other embodiments that second early-minus-latefunction is generally uniformly offset in time or chip spacing from thefirst early-minus-late function.

Under a second procedure, the data processor, carrier tracking module38, or data tracking module 200 determines a second carrier frequencyerror in accordance with a frequency error function (e.g., a secondfrequency error function).

Under a third procedure, the data processor, carrier tracking module 38,or data tracking module 200 determines a second carrier frequency errorin accordance with a dot-product, frequency error function (e.g., asecond frequency error function) that uses prompt QBOC correlation of aprevious epoch or sampling period.

Under a fourth procedure, the data processor, carrier tracking module38, or data tracking module 200 determines a second carrier phase errorin accordance with a phase error function (e.g., second phase errorfunction).

Under a fifth procedure, the data processor, carrier tracking module 38,or data tracking module 200 determines a second carrier phase error inaccordance with a phase error function (e.g., second phase errorfunction) that uses the prompt QBOC correlation.

Under a sixth procedure, a second (carrier) phase error function or thesecond early-minus-late function comprises a QBOC early-minus-latefunction that has a chip spacing of approximately 0.4 chips.

Under a seventh procedure, any of the above procedures of step S714 areexecuted if the primary amplitude is less than the secondary amplitudefor the sampling period.

Step S713 follows step S712, step S714 or step S718. In step S713, thetracking module 200, the measurement generation module 39, or thereceiver 11 processes the determined code error and the determinedcarrier phase error of the received composite signal to track the codeand phase of the received composite signal for estimation of a rangebetween a receiver antenna and satellite transmitter that transmits thereceived composite signal. In an alternate embodiment, the trackingmodule 200, the measurement generation module 39, or the receiver 11processes the determined code error, the determined carrier phase error,and the determined carrier frequency error of the received compositesignal to track the code and phase of the received composite signal forestimation of a range between a receiver antenna and satellitetransmitter that transmits the received composite signal. To determinethe position of the receiver, multiple ranges for multiple carriers fromat least four different satellites are tracked.

Accordingly, FIG. 7B describes DDSel code tracking and DDSel carriertracking, where the carrier tracking includes carrier frequencytracking, carrier phase tracking, or both.

FIG. 7C refers to FIG. 7C-1 and FIG. 7C-2, collectively. FIG. 7C is aflow chart of a method for receiving a composite signal comprises areceiver for receiving a composite signal. Like reference numbers inFIG. 7A, FIG. 7B and FIG. 7C indicate like elements, steps orprocedures.

The method of FIG. 7C, similar to FIG. 7B, uses decision-directedselection (DDSel) to estimate the amplitude of the signal, the carrierfrequency error, and the carrier phase error. However the code errorillustrated in FIG. 7C utilizes MWin-BOC method which linearly combinesthe two EML errors with difference chip spacing.

In FIG. 7C, the multiple window BOC (i.e., MWin-BOC) method is used forpull-in of the code error instead of using the primary amplitude andsecondary amplitude to select correlations for code tracking. Under theMWin-BOC method for pull-in of the code (CD) error, the CD errorestimation results from linear combination of two sets of BOCcorrelations, where the chip spacing for the first BOC correlations isdifferent from that of the secondary BOC correlations and where Equation8 describes the corresponding CD error estimation in greater detail.However, in FIG. 7C, the DDSel method is still used to drive the carriertracking and amplitude estimation.

In FIG. 7C, steps S700, S702, S704, S716, and S706 have been describedgenerally in conjunction with FIG. 7B. Like-designated or like-numberedsteps in FIG. 7B and FIG. 7C indicate like steps or procedures. Becausedifferent steps follow steps S706 and S716 in FIG. 7C than thosefollowing steps set forth in FIG. 7B, FIG. 7C will be describedbeginning with steps S716 and S706.

In the illustrative example of FIG. 7C, step 716 follows step S704. Instep 716, the digital receiver portion 192, the data processor, or acombination of the decision module 35 and the decision augmentationmodule 220 (in FIG. 3A) determines whether (or not) the receiveroperates in a steady state mode. For the received composite BOC signal,if the primary amplitude continuously exceeds the secondary amplitudefor a threshold number of consecutive sampling periods or M epochs, thereceiver (11) or the digital receiver portion 192 switches to or isdetermined to be in the steady state mode, otherwise the receiverremains in or is determined to be in the pull-in mode. The status ofwhether or not the receiver is in the pull-in mode or steady state modemay be determined regularly, or periodically, such as once per epoch orsampling period. In one configuration, the threshold number or M may beany whole number or integer greater than three. However, the thresholdnumber or M may be any suitable number that is determined by empiricalevidence, operational tests, factory setting, programmable setting orotherwise of the receiver that reliably indicates operation of thereceiver in the steady state mode for one or more received channels ofthe composite BOC signal.

The pull-in mode is a pre-alignment state in which the receiver isattempting to align the code, phase and frequency of the local replicasignal with the received composite signal to track and demodulateeffectively, reliably the received composite signal. In the pull-in modethe demodulated energy may be divided between the BOC component and QBOCcomponent of the received signal, making it more difficult toaccurately, reliably recover or decode the modulation on the receivedcomposite signal. In contrast, in the steady state mode, the demodulatedenergy is mostly concentrated in the BOC component.

If the digital receiver portion 192, the data processor, or thecombination of the decision unit 35 and the decision augmentation module220 determine that that the receiver (e.g., one or more channels of thereceived composite signal) is operating in the steady state mode, themethod continues with step S718. However, if the digital receiverportion 192, data processor or the combination of the decision unit 35and the decision augmentation module 220 determine that the receiver isnot operating in the steady state mode, the method continues with stepS706.

In step S718, the digital receiver portion 192, the code tracking module37 or the tracking module 200 processes BOC correlations by determininga code error in accordance with a suitable early-minus-late function forthe steady state mode (e.g., coherent discriminator function). Further,the digital receiver portion, the code tracking module 37 or thetracking module 200 process the BOC correlations by determining acarrier phase error in accordance with a Costas or Phase Locked Loop(PLL) phase error function.

In the steady state mode of step S718, the demodulated energy is mostlyconcentrated in BOC component. Therefore, in the steady state mode thereceiver, data processor, or the decision unit 35 can disregard the QBOCcomponent of the received composite signal under such circumstances,mitigate the noise impact and thus improve the tracking accuracy of thetracking module 200. As shown in FIG. 7B, step S718 only uses the BOCcorrelations, identical to tracking any binary phase shift keying (BPSK)signal, to drive a code tracking module 37, and to drive the carriertracking module 38.

In step S706, a decision unit 35 or electronic data processor determineswhether or not the primary amplitude exceeds (or equals) the secondaryamplitude for a sampling period. For example, in step S706 does theprimary amplitude exceed the secondary amplitude, or is the primaryamplitude greater than or equal to the secondary amplitude? Step S706may be executed regularly or periodically, such as at each epoch or eachsampling period. In step S706, if the primary amplitude exceeds orequals the secondary amplitude, the method continues with step S808.However, if the primary amplitude does not exceed the secondaryamplitude, the method continues with step S810.

In step S808, data processor, a selector, or one or more multiplexers(e.g., second multiplexer 29 for each received channel) select a firstcorrelation or first set of BOC correlations associated with orresulting from: (a) substantially prompt timing (e.g., prompt signal117) of the first signal generator 32 with respect to the receiveddigital signal (102) and (b) a substantially full chip window setting(e.g., established by the code phase window module 157) within thecorrelator module 130.

In step S810, data processor, a selector, or one or more multiplexers(e.g., second multiplexer 29 for each received channel) select a secondcorrelation or second set of QBOC correlations associated with orresulting from: (a) substantially prompt timing (e.g., prompt signal117) of the first signal generator 32 with respect to the receiveddigital signal (102) and (b) a substantially full chip window setting(e.g., established by the code phase window module 157) within thecorrelator module 130.

In the pull-in mode, step S812 follows step S808. In step S812, thedigital receiver portion 192, the data tracking module 200, or thecarrier tracking module 38 process the selected BOC correlations. StepS812 may be carried out by executing one or more procedures that may beapplied separately or cumulatively.

Under a first procedure for carrying out S812, the digital receiverportion 192, the data processor or carrier tracking module 200determines a first carrier frequency error in accordance with afrequency error function (e.g., first frequency error function).

Under a second procedure, the digital receiver portion 192, the dataprocessor or carrier tracking module 200 determines a first carrierfrequency error in accordance with a dot-product, frequency errorfunction (e.g., first frequency error function) using the prompt BOCcorrelation and the prompt BOC correlation of a previous epoch orprevious sampling period.

Under a third procedure, the data processor or carrier tracking module38 determines a first carrier phase error in accordance with a phaseerror function (e.g., first phase error function).

Under a fourth procedure, the data processor or carrier tracking module38 determines a first carrier phase error in accordance with adot-product, phase error function (e.g., first phase error function)using the prompt BOC correlation.

Under a fifth procedure, any of the above procedures or combination ofprocedures of step S812 are executed (e.g., if the primary amplitudeexceeds the secondary amplitude for the sampling period or if theprimary amplitude equals the secondary amplitude for the samplingperiod).

Under the sixth procedure, the first early-minus-late function comprisesa BOC early-minus-late function that has a chip spacing of approximately0.4 chips.

Under the seventh procedure, because the primary amplitude exceeds thesecondary amplitude, in step 812 of FIG. 7C, the data processor,decision unit 35, or digital receiver portion 192 selects only theprompt BOC correlation to generate the BOC-derived carrier frequencyerror and carrier phase error at step 812.

Step S814 follows step S810. In step S814, the digital receiver portion192, the tracking module 200, the code tracking module 37, or thecarrier tracking module 38 process the selected QBOC correlations. StepS814 may be carried out by executing one or more procedures that may beapplied separately or cumulatively.

Under a first procedure for executing step S814, the data processor,carrier tracking module 38, or data tracking module 200 determines asecond carrier frequency error in accordance with a frequency errorfunction (e.g., a second frequency error function).

Under a second procedure, the data processor, carrier tracking module38, or data tracking module 200 determines a second carrier frequencyerror in accordance with a dot-product, frequency error function (e.g.,a second frequency error function) that uses prompt QBOC correlation ofa previous epoch or sampling period.

Under a third procedure, the data processor, carrier tracking module 38,or data tracking module 200 determines a second carrier phase error inaccordance with a phase error function (e.g., second phase errorfunction).

Under a fourth procedure, the data processor, carrier tracking module38, or data tracking module 200 determines a second carrier phase errorin accordance with a phase error function (e.g., second phase errorfunction) that uses the prompt QBOC correlation.

Under a fifth procedure, any of the above procedures or combination ofprocedures of step S814 are executed (e.g., if the primary amplitude isless than the secondary amplitude for the sampling period).

Under a sixth procedure, the second early-minus-late function comprisesa QBOC early-minus-late function that has a chip spacing ofapproximately 0.4 chips.

Under the seventh procedure, because the primary amplitude does notexceed the secondary amplitude, in step 814 of FIG. 7C, the dataprocessor, decision unit 35, or digital receiver portion 192 selectsonly the prompt QBOC correlation to generate the QBOC-derived carrierfrequency error and carrier phase error at step 814.

In step S715, the data processor or digital receiver portion 192: (a)selects a first set of BOC correlations with first chip spacing (e.g.,approximately 0.25 chips) to form a first code error (e.g., first EMLcode error); (b) selects a second set of BOC correlations with a secondchip spacing (e.g., approximately 0.125 chips), distinct from the firstchip spacing, to form a second code error (e.g., second EML code error),and (c) linearly combines the first code error and the second code errorto form a third code error or third code error estimation. Step S715 mayuse Equation 8, as previously set forth in this document.

Step S711 follows step S715 or step 718. In step S711, the trackingmodule 200, the measurement generation module 39, or the receiver 11processes the selected correlations to track the code and carrier of thereceived composite signal for estimation of a range between a receiverantenna and satellite transmitter that transmits the received compositesignal.

FIG. 7D refers to FIG. 7D-1 and FIG. 7D-2 collectively. Like referencenumbers in FIG. 7C and FIG. 7D indicate like steps or procedures. FIG.7C describes use of MWin-BOC for code tracking and the use of DDSel forcarrier tracking. FIG. 7D is similar to FIG. 7C, except that FIG. 7Dsubstitutes S708 and S710 for steps S808 and S810 of FIG. 7C,respectively.

In step S708, data processor, a selector, or one or more multiplexers(e.g., second multiplexer 29 for each received channel) select a set ofBOC correlations in accordance with a BOC correlation function for thesampling period if the primary amplitude exceeds or equals the secondaryamplitude for the sampling period. For example, the second multiplexer29 selects a set of BOC correlations (e.g., via the BOC selection signal123 provided to multiplexer 29) in accordance with a BOC correlationfunction for the sampling period if the primary amplitude exceeds orequals the secondary amplitude for the sampling period.

In step S710, if the primary amplitude is less than the secondaryamplitude or if the secondary amplitude exceeds the primary amplitude,the data processor, selector, or one or more multiplexors (e.g., secondmultiplexer 29 for each received channel) select a set of QBOCcorrelations in accordance with a QBOC correlation function for currentsampling period to generate the error estimations to drive the code (CD)and carrier (CR) feedback loop. In one embodiment, the zero crossingpoint for the QBOC correlation function is offset by approximately

$\frac{1}{4N_{BOC}}$against the zero crossing point for the BOC correlation function.

FIG. 8 is a graph of a BOC correlation function (e.g., for a BOC(1,1)signal) and QBOC correlation function (e.g., for a QBOC(1,1) signal)using a closed-formula representation. In FIG. 8, the vertical axisprovides the correlation output or amplitude and the horizontal axis isin time (e.g., chips). The BOC correlation function (e.g., for aBOC(1,1) signal) is shown by solid line 701, whereas the QBOCcorrelation function (e.g., for a QBOC(1,1) signal) is shown by dashedline 702.

FIG. 9 is a graph of decision-directed selection (DDsel) amplitudederived from either the BOC or the QBOC signal components, and isrepresentative of an aggregate correlation function (of BOC and QBOCcorrelations over multiple sampling periods without simultaneous use ofBOC and QBOC correlations for any received channel during any singlesampling period or any single epoch). An aggregate correlation functionmeans a dual-correlation function (of BOC and QBOC correlations overmultiple sampling periods, without simultaneous use of BOC and QBOCcorrelations for any received channel during any single sampling periodor epoch), a decision-directed correlation function (of BOC and QBOCcorrelations over multiple sampling periods, without simultaneousselection of BOC and QBOC correlations for any received channel)), orintermittently alternating correlation functions over multiple samplingperiods resulting from either BOC or QBOC correlation at each samplingperiod or epoch. The vertical axis represents amplitude, whereas thehorizontal axis represents time in chips. For example, zero on the timeaxis represents a perfectly correlated BOC signal with the local replicaof the BOC signal at the receiver during one or more sampling intervals,for example.

FIG. 10 illustrates a method of selecting or identifying a pull-in modeor a steady state mode for operation of any receiver, system or methoddescribed in this document. The method of FIG. 10 begins in step S701.

In step S701, a receiver is initialized, turned on or powered up and hasthe pull-in mode as the default or starting mode of operation. Thepull-in mode is a pre-alignment state in which the receiver isattempting to align the code, phase and frequency of the local replicasignal with the received composite signal to track and demodulateeffectively, reliably the received composite signal. In the pull-in modethe demodulated energy may be divided between the BOC component and QBOCcomponent of the received signal, making it more difficult toaccurately, reliably recover or decode the modulation on the receivedcomposite signal. In contrast, in the steady state mode, the demodulatedenergy is mostly concentrated in the BOC component.

In step S717, a data processor or a digital receiver portion 192determines if a counter value (e.g., counter A) or register (e.g., of adecision augmentation module 220 or counter) is greater than or equal toM. For the received composite BOC signal, if the primary amplitudecontinuously exceeds the secondary amplitude for a threshold number ofconsecutive sampling periods or M epochs, the receiver (11) or thedigital receiver portion 192 switches to or is determined to be in thesteady state mode, otherwise the receiver remains in or is determined tobe in the pull-in mode. In one configuration, the threshold number or Mmay be any whole number or integer greater than three. However, thethreshold number or M may be any suitable number that is determined byempirical evidence, operational tests, factory setting, programmablesetting or otherwise of the receiver that reliably indicates operationof the receiver in the steady state mode for one or more receivedchannels of the composite BOC signal. If the counter value is greaterthan or equal to M, the method continues with step S724. However, if thecounter value is not greater than or equal to M, the method continueswith step S702.

In step S702, a first detector 201 (e.g., in FIG. 1, FIG. 3 and FIG. 4)or data processor detects a primary amplitude of the BOC component. StepS702 may be executed in accordance with various techniques, which may beapplied alternately or cumulatively.

Under a first technique, the first detector 201 or data processordetects or measures the signal energy or aggregate power of the BOCcomponent comprising in-phase BOC component and a quadrature-phase BOCcomponent.

Under a second technique, the first detector 201 or data processorapplies the correlation P_(BOC) ^((0,1)) of Equation 3 to provide anamplitude (e.g., an ideal amplitude) estimate for a BOC component of thereceived composite signal during a sampling period or a time interval.For example, the first detector 201 applies a correlation P_(BOC)^((0,1)) of Equation 3 to provide ideal amplitude estimation for BOCcomponent: Amp(Y)=√{square root over (I_Y²+Q_Y²)},

where

I_Y is the in-phase BOC component, and

Q_Y is the quadrature-phase BOC component.

Under a third technique, the first detector 201 or the data processoruses the linear approximation Equation 4 to simplify or replace thecalculation of Equation 3. In one embodiment, the linear approximationof Equation 4 has a bias that is acceptable and facilitates rapidestimation of the primary amplitude. Under the third technique, theestimating or detecting of the primary amplitude or the secondaryamplitude (e.g., by detectors 201, 211) for the sampling period is basedon the following equations that use linear approximation separately forthe BOC and QBOC signal components:Amp^(approx)(Y)

max(|I_Y|,|Q_Y|)+μmin(|I_Y|,|Q_Y|),where

Y=P_(BOC) ^(W) ^(X) ,

μ is a selected scale or a constant scaling factor (e.g., 0.5),

W_(X) is the window size of the correlator,

I_Y is the in-phase xBOC component, and

Q_Y is the quadrature-phase xBOC component,

xBOC refers to the BOC or QBOC signal component of the receivedcomposite signal. In the above equation, for instance, the firstdetector 201 detects the BOC signal component and substitutes the BOCsignal component for xBOC.

In step S704, a second detector 211 (in FIG. 1, FIG. 3 and FIG. 4) orthe data processor detects a secondary amplitude of the QBOC component.Step S704 may be executed in accordance with various techniques, whichmay be applied alternately or cumulatively.

Under a first technique, the second detector 211 or data processordetects or measures the signal energy or aggregate power of the QBOCcomponent comprising in-phase QBOC component and a quadrature-phase QBOCcomponent.

Under a second technique, the second detector 211 or data processorapplies a correlation P_(BOC) ^((0,1)) of Equation 3 to provide idealamplitude estimation for QBOC component: Amp(Y)=√{square root over(I_Y²+Q_Y²)}, where

I_Y is the in-phase QBOC component, and

Q_Y is the quadrature-phase QBOC component.

Under a third technique, the linear approximation Equation 4 simplifiesor replaces the calculation of FIG. 3. In one embodiment, the linearapproximation of Equation 4 has a bias that is acceptable andfacilitates rapid estimation of the primary amplitude. Under the thirdtechnique, the estimating or detecting the secondary amplitude (e.g., bythe second detector 211 or data processor) for the sampling period isbased on the following equations that use linear approximationseparately for the QBOC signal components:Amp^(approx)(Y)

max(|I_Y|,|Q_Y|)+μmin(|I_Y|,|Q_Y|),where

Y=P_(BOC) ^(W) ^(X) ,

μ is a selected scale or a constant scaling factor (e.g., 0.5),

W_(X) is the window size of the correlator,

I_Y is the in-phase QBOC component, and

Q_Y is the quadrature-phase QBOC component.

In step S706, a decision unit 35 or electronic data processor determineswhether or not the primary amplitude is greater than or equal to thesecondary amplitude for a sampling period. For example, in step S706does the primary amplitude exceed the secondary amplitude? In step S706,if the primary amplitude is greater than or equal the secondaryamplitude, the method continues with step S720. However, if the primaryamplitude in not greater than or equal to the secondary amplitude, themethod continues with step S722.

In step S722, the data processor or digital receiver portion 192 resetsthe counter (e.g., counter A), register or data storage deviceassociated with the decision augmentation module 220 and then returns tostep S717. For example, the data processor or digital receiver portion192 sets the counter (e.g., counter A), register or the data storagedevice to zero.

In step S720, the data processor or digital receiver portion 192increments the counter (e.g., counter A), a register or data storagedevice associated with the decision augmentation module 220. Forexample, the data processor or digital receiver portion 192 incrementsby one the counter (e.g., counter A), a register or data storage deviceassociated with the decision augmentation module 220.

The methods of FIG. 11 through FIG. 14, inclusive, may incorporate byreference techniques, processes, features or elements from any othermethods or systems disclosed in this document. For example, any methodin FIG. 11 through FIG. 14, inclusive, may incorporate one or moreembodiments or variations of any similar steps disclosed in FIG. 7Athrough FIG. 10 and the accompanying text. Like reference numbers shallrefer to like elements in all drawings. As the context requires, liketerminology can refer to like elements or features in the drawings.

FIG. 11 discloses a method of receiving a composite signal. The methodof FIG. 11 is similar to the method of FIG. 7A, except the method ofFIG. 11 replaces step S711 with S751. Like reference numbers in FIG. 7Aand FIG. 11 indicate like steps or procedures.

Step S751 may be executed after step S708 or S710. In step S751, thedata processor or digital receiver portion 192 processes the selectedcorrelations (e.g., selected substantially prompt BOC correlations orsubstantially prompt QBOC correlations) to track a carrier of thereceived composite signal for estimation of a range between a receiverantenna and a satellite transmitter that transmits the receivedcomposite signal.

FIG. 12 discloses a method of receiving a composite signal. FIG. 12refers to FIG. 12-1 and FIG. 12-2 collectively. The method of FIG. 12 issimilar to the method of FIG. 7A, except the method of FIG. 12 replacesstep S711 with step S811. Step S811 comprises steps S752, S753 and S754.Like reference numbers in FIG. 7A and FIG. 12 indicate like steps orprocedures.

In step S752, the data processor or digital receiver portion 192processes the selected correlations (e.g., selected substantially promptBOC correlations or substantially prompt QBOC correlations) during thesampling period, or both correlations (e.g., substantially prompt BOCand QBOC correlations) over successive sampling periods, to track acarrier of the received composite signal. For example, the dataprocessor or digital receiver portion 192 processes a first subset ofthe selected set of correlations to track a carrier of the receivedcomposite signal for estimation of a range between a receiver antennaand satellite transmitter that transmits the received composite signal.In one embodiment, the first subset comprises the prompt BOC correlationor the prompt QBOC correlation.

In step S753, the data processor or digital receiver portion 192processes the selected correlations (e.g., selected substantially earlyand late BOC correlations, or substantially early and late QBOCcorrelations) during the sampling period, or both correlations (e.g.,substantially early and late BOC correlations and substantially earlyand late QBOC correlations) over successive sampling periods, to track acode of the received composite signal. For example, the data processoror digital receiver portion 192 processes a second subset of theselected correlations during the sampling period. The second subsetcomprises a BOC pair of an early BOC correlation and a late BOCcorrelation, or QBOC pair of a respective early QBOC correlation and arespective late QBOC correlation, where each of the pairs of thecorrelations has a first chip spacing to drive the code tracking. Thedata processor or digital receiver portion 192 is capable of processingthe second subset of the selected set of correlations to form a firstcode error.

In step S754, the data processor or digital receiver portion 192estimates a range between a receiver antenna and satellite transmitterthat transmits the received composite signal based on the trackedcarrier, the tracked code, or both.

As indicated by the dashed lines, steps S752, S753 and S754 may becollectively labeled as step S811. In certain embodiments, step S811 issomewhat similar to step S711 of FIG. 7A and aspects or features of S711may be applied to step S811.

FIG. 13 discloses a method of receiving a composite signal. FIG. 13refers to FIG. 13-1 and FIG. 13-2, collectively. The method of FIG. 13is similar to the method of FIG. 7A, except the method of FIG. 13replaces step S711 with steps S752, S755, S756, and S754. Like referencenumbers in FIG. 7A, FIG. 12 and FIG. 13 indicate like steps orprocedures.

In step S752, the data processor or digital receiver portion 192processes the selected correlations (e.g., selected substantially promptBOC correlations or substantially prompt QBOC correlations) during thesampling period, or both correlations (e.g., substantially prompt BOCand QBOC correlations) over successive sampling periods, to track acarrier of the received composite signal.

In step S755, the data processor or digital receiver portion 192 forms afirst code error using a set of BOC correlations with a first chipspacing and a second code error using another set of BOC correlationswith a second chip spacing, where the first and the second chip spacingsare different. In one embodiment, the first chip spacing is within arange of approximately 0.25 chips to approximately 0.5 chips. In oneembodiment, the second chip spacing is within a range of approximately0.125 chips to approximately 0.25 chips.

In step S756, the data processor or the digital receiver portion 192combines the first code error and the second code error (e.g., into athird code error) to drive the code tracking.

In step S754, the data processor or digital receiver portion 192estimates a range between a receiver antenna and satellite transmitterthat transmits the received composite signal based on the trackedcarrier, the tracked code, or both.

FIG. 14 discloses a method of receiving a composite signal. The methodof FIG. 14 starts in step S700. Like references in FIG. 14 and otherdrawings indicate like elements, steps or procedures.

In step S700, a receiver 11 (e.g., satellite navigation receiver) or adigital receiver portion 192 receives a binary offset carrier (BOC)modulated signals to extract a BOC component by mixing or combining witha local BOC replica, and to derive a quadrature BOC (QBOC) component bycombining with a local QBOC replica. In one embodiment, the BOCcomponent comprises an in-phase BOC component and a quadrature-phase BOCcomponent and the QBOC component comprises in-phase QBOC component and aquadrature-phase QBOC component. For example, the receiver 11 comprisesa code mixer 42 (FIG. 1) for mixing or combining the BOC signal with thelocal BOC or QBOC replica from one or more signal generators (e.g.,multiplexed output of a first signal generator 32 and second signalgenerator 31).

In step S702, a first detector 201 (e.g., in FIG. 1, FIG. 3 and FIG. 4)or data processor detects a primary amplitude of the BOC componentduring a sampling period or interval. Step S702 may be executed inaccordance with various techniques, which may be applied alternately orcumulatively, as more fully described in conjunction with FIG. 7A.

In step S716, the data processor or digital receiver portion 192determines if a receiver is in a steady-state mode. Step S716 may beexecuted in accordance with various techniques, which may be appliedalternatively or cumulatively.

Under a first technique, the data processor or digital receiver portion192 determines that the receiver is operating in a steady-state mode ifthe detected primary amplitude equals or exceeds a threshold amplitudevalue over one or more sampling periods (e.g., over one or more previoussampling periods prior to a current sampling period.) For instance, thethreshold amplitude value may be based on the primary amplitude orsignal strength of the received signal that indicates a majority or mostof the energy of the received signal lies within the BOC componentversus the QBOC component for a sampling period.

Under a second technique, the data processor or digital receiver portion192 determines that receiver is operating in the steady-state mode bydetermining that the primary amplitude is continuously greater than thesecondary amplitude for equal to or greater than a threshold duration,wherein the steady-state mode is mutually exclusive to a pull-in mode.The threshold duration may equal one or more sampling periods or epochsor may comprise a generally continuous duration of sampling periods orepochs in which the primary amplitude is greater than the secondaryamplitude.

In step S716 if the receiver is in a steady-state mode, the methodcontinues with step S708. However, if the receiver is not in thesteady-state mode or is in the pull-in mode, the method continues withstep S801.

In step S708, the data processor or digital receiver portion 192 selectsa set of BOC correlations (e.g., in accordance with a BOC correlationfunction) for a sampling period (e.g., current sampling period).

In step S801, the data processor or digital receiver portion 192 selectsBOC or QBOC correlations that correspond to the signal component of thereceived composite signal with the greatest amplitude during a samplingperiod or with a greatest amplitude (e.g., mean or average amplitude)over one or more previous sampling periods.

In step S802, the data processor or digital receiver portion 192processes the selected correlations (e.g., BOC correlations for steadystate mode, or BOC or QBOC correlations for pull-in mode) during thesampling period to track a carrier of the received composite signal.

In step S803, the data processor or digital receiver portion 192 forms afirst code error using the selected correlations (e.g., BOC correlationsfor steady-state mode, or BOC or QBOC correlations for pull-in mode)with a first chip spacing to drive the code tracking.

In step S754, the data processor or digital receiver portion 192estimates a range between a receiver antenna and satellite transmitterthat transmits the received composite signal based on the trackedcarrier, the tracked code, or both.

The method and system disclosed in this document uses adecision-directed selection (DDSel). Under the decision-directedselection, the method and system of this disclosure uses the either BOCor QBOC terms, whichever component retains more signal energy to processboth code and carrier acquisition or pull in, instead of using both BOCand QBOC. Because the method and system of this disclosure uses decisiondirected selection (DDSel), the method and system is well suited forreducing computational load as it either processing BOC or QBOC termsbased on the selection, as opposed to processing both BOC and QBOCterms. In the method and system disclosed in this document, DDSel canpromptly, reliably select the proper BOC or QBOC processing to acquireor pull-in the BOC signal at the receiver, where a wide correlator canemploy a half-chip spacing delay line during pull-in or acquisitionbecause of the absence of zero crossing points in the aggregatecorrelation function. Further, DDSel supports unambiguous determinationof frequency error and phase error by eliminating or reducing theirdependency on the code error through using dot-product carrier errorfunction. DDSel uses an unambiguous DDSel frequency error term forcarrier pull in or acquisition of the received signal. The method andsystem is well suited for providing faster pull-in performance thancertain prior art because uses the error component (BOC or QBOC errorcomponent) with better signal-to-noise ratio (SNR), which helps toreduce the loop measurement fluctuation resulting from the noise.

For the input of BOC(m,n) or QBOC(m,n), the DDSel technique can applywithin context of a wide correlator, a narrow correlator, a windowcorrelator, or an combination of the foregoing correlators. DDSel canconstrain the window/chip spacing to pull in or acquire the code of thereceived signal. DDSel uses envelope amplitude estimation to determinehow to process the received signal in accordance with alternate BOC orQBOC processing paths. The method and system disclosed in this documentcan simplify the amplitude estimation with an approximation equation, aspreviously explained. In certain configurations, such as with the use ofthe above approximation equation, DDSel is capable of providingamplitude detection speed that is commensurate with a received BOCsignal input applied to a half-chip spacing delay line.

An alternate embodiment of a method for receiving the composite signalcombines two BOC early-minus-late code errors with different chipspacing to drive the code tracking. Under such an alternate operatingmode, however, DDSel still provides the carrier tracking and amplitudeestimation.

In another alternate embodiment, receiver operation mode can be switchedfrom DDSel for acquisition mode, to single-window BOC (SWin-BOC) forlock in or post acquisition mode.

Having described the preferred embodiment, it will become apparent thatvarious modifications can be made without departing from the scope ofthe invention as defined in the accompanying claims.

The following is claimed:
 1. A method for receiving a received compositesignal, the method comprising: receiving a plurality of compositesignals from corresponding satellites, the composite signal, from eachsatellite, comprising binary offset carrier (BOC) modulated signals thatare processed to obtain a BOC component and to derive a quadrature BOC(QBOC) component; detecting a primary amplitude of the BOC component ofone of the received composite signals based on correlation data;detecting a secondary amplitude of the QBOC component of the one of thereceived composite signals based on the correlation data; determiningwhether the primary amplitude exceeds the secondary amplitude for asampling period; selecting a set of BOC correlations in accordance witha BOC correlation function for the sampling period if the primaryamplitude exceeds the secondary amplitude for the sampling period; andselecting a set of QBOC correlations in accordance with a QBOCcorrelation function for the sampling period if the secondary amplitudeexceeds the primary amplitude for the sampling period; and processingthe selected correlations to track a carrier of the one of the receivedcomposite signals for estimation of a range between a receiver antennaand the corresponding satellite transmitter that transmits the one ofthe received composite signals.
 2. The method according to claim 1wherein the processing of the selected correlations further comprisesusing the BOC correlations for: determining a first code error inaccordance with a first early-minus-late function; determining a firstcarrier frequency error in accordance with a frequency error function;and determining a first carrier phase error in accordance with a phaseerror function, if the primary amplitude exceeds the secondary amplitudefor the sampling period.
 3. The method according to claim 2 wherein thefirst early-minus-late function comprises a BOC early-minus-latefunction that has a chip spacing of approximately 0.4 chips.
 4. Themethod according to claim 1 wherein the processing of the selectedcorrelations further comprises using the set of QBOC correlations for:determining a second code error in accordance with a secondearly-minus-late function; determining a second carrier frequency errorin accordance with a frequency error function; and determining a secondcarrier phase error in accordance with a phase error function, if theprimary amplitude is less than the secondary amplitude for the samplingperiod.
 5. The method according to claim 4 wherein the secondearly-minus-late function comprises a QBOC early-minus-late functionthat has a chip spacing of approximately 0.4 chips.
 6. The methodaccording to claim 1 further comprising: executing the above selecting,detecting and determining steps during a pull-in mode; identifying asteady state mode of a receiver after pull-in or acquisition of the codeand phase of the one of the received composite signals; and selectingonly the set of BOC correlations for the steady state mode and usingonly a BOC error function during the steady state mode for an epoch,unless a cycle slip occurs.
 7. The method according to claim 1 whereinthe set of BOC correlations is modeled by the following equation to forma first error function or first-early minus late function:$\begin{matrix}{{TAP}_{BOC}^{w_{X}} = {{AW}_{X}{\sum\limits_{{i = 0},{{di} = {F_{Chip}/F_{S}}}}^{N_{Chip}}\left\{ \begin{matrix}\begin{matrix}{c_{i}{c_{i - \tau}.}} \\{{\sin\left( {2\pi\; N_{BOC}i} \right)}{{\sin\left( {2\pi\;{N_{BOC}\left( {i - \tau} \right)}} \right)}.}}\end{matrix} \\{\mathbb{e}}^{j\;\delta\;\theta_{i}}\end{matrix} \right.}}} \\{= {\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}{\mathbb{e}}^{j\delta\varphi}}} \\{{= {\overset{\overset{{I\_ TAP}_{BOC}^{W_{X}}}{︷}}{\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}\cos\;{\delta\varphi}} + {j\underset{\underset{{Q\_ TAP}_{BOC}^{W_{X}}}{⎴}}{\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}\sin\;\delta\;\varphi}}}},}\end{matrix}$ where TAP_(BOC) ^(W) ^(x) , is the correlation based onthe BOC input and a BOC local replica with a selection of pseudo-noisecode (PN) TAP, where PN TAP can be early, prompt or late, i is chipphase in unit of chips, N _(Chip) is number of chips per millisecond, diis sample phase increment in unit of chips, A is the amplitude of signalat level of millisecond integration, W_(x) is the window size in chips,N_(BOC) is equal to m/n of BOC (m,n)where m is f_(m)/f_(c)and n isf_(n)/f_(c),f_(m) , is a first subcarrier frequency, f_(n) is the actualchip frequency, and f_(c) is the reference chipping rate, F_(Chip) isthe chipping rate in chips/second, F_(S) is the sampling rate insamples/second, R(τ) is the correlation function for PN code, τ is thechip phase error estimated at receiver side, δθ_(i) is the carrier phaseestimation error for each sample, δφ is the average carrier phaseestimation error over N_(Chip) period, I_TAP_(BOC) ^(W) ^(x) is thein-phase or reel part of TAP_(BOC) ^(W) ^(x) , and Q_TAP_(BOC) ^(W)isthe quadrature or imaginary part of TAD_(BOC) ^(W) ^(x) ; and c is thecode.
 8. The method according to claim 1 wherein the set of QBOCcorrelations is modeled by the following equation to form a secondearly-minus-late function: $\begin{matrix}{{TAP}_{QBOC}^{w_{X}} = {{AW}_{X}{\sum\limits_{{i = 0},{{di} = {F_{Chip}/F_{S}}}}^{N_{Chip}}\left\{ \begin{matrix}\begin{matrix}{c_{i}{c_{i - \tau}.}} \\{{\sin\left( {2\pi\; N_{BOC}i} \right)}{{\cos\left( {2\pi\;{N_{BOC}\left( {i - \tau} \right)}} \right)}.}}\end{matrix} \\{\mathbb{e}}^{j\;\delta\;\theta_{i}}\end{matrix} \right.}}} \\{= {\frac{A}{2}W_{X}{R(\tau)}{\sin\left( {2\pi\; N_{BOC}\tau} \right)}{\mathbb{e}}^{j\delta\varphi}}} \\{{= {\overset{\overset{{I\_ TAP}_{QBOC}^{W_{X}}}{︷}}{\frac{A}{2}W_{X}{R(\tau)}{\sin\left( {2\pi\; N_{BOC}\tau} \right)}\cos\;{\delta\varphi}} + {j\underset{\underset{{Q\_ TAP}_{QBOC}^{W_{X}}}{⎴}}{\frac{A}{2}W_{X}{R(\tau)}{\sin\left( {2\pi\; N_{BOC}\tau} \right)}\sin\;\delta\;\varphi}}}},}\end{matrix}$ where TAP_(QBOC) ^(W) ^(X) is the correlation based on theBOC input and a QBOC local replica with a selection of pseudo-noise code(PN) TAP, where PN TAP can be early, prompt or late, i is chip phase inunit of chips, N_(Chip) is number of chips per millisecond, di is samplephase increment in unit of chips, A is the amplitude of signal at levelof millisecond integration, W_(X) is the window size in chips, N_(BOC)is equal to m/n of BOC (m,n),where m is f_(m)/f_(c),and n is f_(n)/f_(c),f_(m) is a first subcarrier frequency, f_(n) is the actual chipfrequency, and f_(c) is the reference chipping rate, F_(chip) is thechipping rate in chips/second, F_(S) is the sampling rate insamples/second, R(τ) is the correlation function for PN code, τ is thechip phase error estimated at receiver side, δθ_(i) is the carrier phaseestimation error for each sample, δφ is the average carrier phaseestimation error over N_(Chip) period, I_TAP_(QBOC) ^(w) ^(x) is thein-phase or real part of TAP_(QBOC) ^(W) ^(x) , and Q_TAP_(QBOC) ^(w)^(xw) is the quadrature or imaginary part of TAP_(QBOC) ^(W) ^(X) ; andc is the code.
 9. The method according to claim 1 wherein the detectingof the primary amplitude or the secondary amplitude for the samplingperiod is based on the following equations that use linear approximationseparately for the BOC and QBOC signal components:Amp^(Approx)(Y)

max(|I_Y|,|Q_Y|)+μmin(|I_Y|,|Q_Y|), where Y=TAP_(xBOC) ^(W) ^(X) , acorrelation based on the xBOC component and a xBOC local signal with aselection of the pseudo-noise code (PN) TAP, where the PN TAP can beearly, prompt or late; μ is a selected scale or constant scaling factor,W_(x)is a window size of a correlator, I_Y is an in-phase xBOCcomponent, and Q_Y is a quadrature-phase xBOC component, xBOC refers tothe BOC or QBOC signal component of the one of the received compositesignals.
 10. The method according to claim 1 wherein the primaryamplitude comprises the combined signal power of a BOC in-phasecomponent and pen a BOC quadrature phase component of the BOC component;and wherein the secondary amplitude comprises the combined signal powerof a QBOC in-phase component and a QBOC quadrature-phase component ofthe QBOC component.
 11. The method according to claim 1 wherein a zerocrossing point for a BOC amplitude function offsets by approximately$\frac{1}{4N_{BOC}}$ against a zero crossing point for a QBOC amplitudefunction, where N _(BOC) is equal to m/n of BOC (m,n),where m isf_(m)/f_(c) and n is f_(n)/f_(c), f_(m) is a first subcarrier frequency,f_(n), is the actual chip frequency, and f_(c) is the reference chippingrate.
 12. The method according to claim 1 further comprising: for eachsuccessive sampling period in a code loop, selecting the set of BOCcorrelations or the set of QBOC correlations to form theearly-minus-late estimation of a code error based on the greateramplitude resulting from the BOC component or from the QBOC component,respectively, which is modeled by the following equation:${\tau = \frac{{E_{xBOC}} - {L_{xBOC}}}{A_{DDSel}}},$ where xBQC iseither BOC or QBOC, E_(xBOC) corresponds toP_(xBOC) ^((0,Wa)), L_(xBOC)corresponds to P_(xBOC) ^((1−Wa,1)), and where W_(a) is a window size ofa correlator, and where P_(xBOC) ^(O,Wa)), is an amplitude of thereceived xBOC signal with one window size of the correlator and whereP_(xBOC) ^(1−Wa,1)), is an amplitude of the received xBOC signal withanother window size of the correlator.
 13. The method according to claim1 further comprising: for each successive sampling period in a carrierphase loop, the selected set of xBOC correlations, with greateramplitude, form a carrier phase error estimation, wherein across-product using an in-phase and quadrature phase component resolvesthe ambiguity from sine or cosine of BOC misalignment, where xBOC meansBOC or QBOC.
 14. The method according to claim 1 wherein: selecting theset of BOC correlations in accordance with the BOC correlation functionfor the sampling period, and combining with the set of BOC correlationsfor a previous sampling period to form a BOC cross-product carrierfrequency estimation if the primary amplitude exceeds or equals thesecondary amplitude for the sampling period, or selecting the set ofQBOC correlations in accordance with the QBOC correlation function forthe sampling period, and combining with the set of QBOC correlations forthe previous sampling period to form a QBOC cross-product carrierfrequency estimation if the secondary amplitude exceeds the primaryamplitude.
 15. The method according to claim 1 further comprisingforming a first code error using a set of BOC correlations with a firstchip spacing and a second code error using another set of BOCcorrelations with a second chip spacing, and combining the first codeerror and the second code error to drive a code tracking in accordancewith the following composite error function:$\tau_{{MWin} - {BOC}} = {{\alpha\frac{{P_{BOC}^{({0,{Wa}})}} - {P_{BOC}^{({{1 - {Wa}},1})}}}{A_{DDSel}}} + {\beta\frac{{P_{BOC}^{({0,{Wb}})}} - {P_{BOC}^{({{1 - {Wb}},1})}}}{A_{DDSel}}}}$where α is the linear gain applied on the first BOC-derived code errorestimation with window Wa, β is the linear gain applied on the secondaryBOC-derived code error estimation with window Wb. and P_(xBOC)^((O,Wa)),is an amplitude of the received BOC signal with one windowsize of a correlator and where P_(xBOC) ^((1−Wa,1)), is an amplitude ofthe received BOC signal with another window size of the correlator. 16.The method according to claim 1 further comprising: after codeacquisition of the received signal and during a code lock mode, only usethe set of BOC correlations to drive a code and carrier tracking. 17.The method according to claim 1 wherein the processing the selectedcorrelations further comprises tracking the carrier phase and frequencyof the carrier during a pull-in mode.
 18. The method according to claim1 wherein the processing the selected correlations further comprisestracking the code, carrier phase, and carrier frequency of the receivedcomposite signal during a pull-in mode.
 19. A system for receiving areceived composite signal, the system comprising: a receiver front endfor receiving a composite signal comprising binary offset carrier (BOC)modulated signals; a first signal generator for generating a chipreplica for any sampling period; a second signal generator forgenerating a local BOC signal or a quadrature BOC signal for anysampling period; an electronic data processor adapted to obtain a BOCcomponent via the local BOC signal and to derive a quadrature BOC (QBOC)component; a first detector for detecting a primary amplitude of the BOCcomponent of the received composite signal based on correlation data; asecond detector for detecting a secondary amplitude of the QBOCcomponent of the received composite signal based on the correlationdata; an evaluator of the data processor determining whether the primaryamplitude exceeds the secondary amplitude for a sampling period; aselector of the data processor for selecting a set of BOC correlationsin accordance with a BOC correlation function for the sampling period ifthe primary amplitude exceeds the secondary amplitude for the samplingperiod; and the selector is adapted to select a set of QBOC correlationsin accordance with a QBOC correlation function for the sampling periodif the secondary amplitude exceeds the primary amplitude for thesampling period wherein the selected xBOC correlations with greateramplitude supports unambiguous code acquisition of the received signal;where xBOC means BOC or QBOC; and a tracking module of the electronicdata processor for processing the selected correlations to track acarrier of the received composite signal for estimation of a rangebetween a receiver antenna and each satellite transmitter that transmitsthe received composite signal.
 20. The system according to claim 19wherein the tracking module is adapted to use the BOC correlations for:determining a first code error in accordance with a firstearly-minus-late function; determining a first carrier frequency errorin accordance with a frequency error function; and determining a firstcarrier phase error in accordance with a phase error function, if theprimary amplitude exceeds the secondary amplitude for the samplingperiod.
 21. The system according to claim 20 wherein the firstearly-minus-late function comprises a BOC early-minus-late function thathas a chip spacing of approximately 0.4 chips.
 22. The system accordingto claim 19 wherein the tracking module is adapted to use the set ofQBOC correlations for: determining a second code error in accordancewith a second early-minus-late function; determining a second carrierfrequency error in accordance with a frequency error function; anddetermining a second carrier phase error in accordance with a phaseerror function, if the primary amplitude is less than the secondaryamplitude for the sampling period.
 23. The system according to claim 22wherein the second early-minus-late function comprises a QBOCearly-minus-late function that has a chip spacing of approximately 0.4chips.
 24. The system according to claim 19 further comprising: acounter for identifying a steady state mode of a receiver after pull-inor acquisition of a code and phase of the received composite signal ifthe primary amplitude is greater than a second amplitude for a thresholdnumber of successive sampling periods or epochs; and a selector forselecting only the set of BOC correlations for the steady state mode andusing only a BOC error function during the steady state mode for anepoch, unless a cycle slip occurs.
 25. The system according to claim 19wherein the set of BOC correlations is modeled by the following equationto form a first-early minus late function: $\begin{matrix}{{TAP}_{BOC}^{w_{X}} = {{AW}_{X}{\sum\limits_{{i = 0},{{di} = {F_{Chip}/F_{S}}}}^{N_{Chip}}\left\{ \begin{matrix}\begin{matrix}{c_{i}{c_{i - \tau}.}} \\{{\sin\left( {2\pi\; N_{BOC}i} \right)}{{\sin\left( {2\pi\;{N_{BOC}\left( {i - \tau} \right)}} \right)}.}}\end{matrix} \\{\mathbb{e}}^{{j\theta}_{i}}\end{matrix} \right.}}} \\{= {\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}{\mathbb{e}}^{j\delta\varphi}}} \\{{= {\overset{\overset{{I\_ TAP}_{BOC}^{W_{X}}}{︷}}{\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}\cos\;{\delta\varphi}} + {j\underset{\underset{{Q\_ TAP}_{BOC}^{W_{X}}}{⎴}}{\frac{A}{2}W_{X}{R(\tau)}{\cos\left( {2\pi\; N_{BOC}\tau} \right)}\cos\;\delta\;\varphi}}}},}\end{matrix}$ where TAP_(BOC) ^(W) ^(X) is the correlation based on theBOC input and the BOC local replica with a selection of pseudo-noisecode (PN) TAP, where PN TAP can be early, prompt or late, i is chipphase in unit of chips, N_(Chip) is number of chips per millisecond, diis sample phase increment in unit of chips, A is the amplitude of signalat level of millisecond integration, W_(x) is the window size in chips,N_(BOC) is equal to in of BOC (m,n),where m is f_(m)/f_(c)and n isf_(n)/f_(c),f_(m) is a first subcarrier frequency, f_(n) is the actualchip frequency, and f_(c) is the reference chipping rate, F_(chip) isthe chipping rate in chips/second, F_(S) is the sampling rate insamples/second, R(τ) is the correlation function for PN code, τ is thechip phase error estimated at receiver side, δθ_(i) is the carrier phaseestimation error for each sample, δφ is the average carrier phaseestimation error over N_(Chip) period, I_TAP_(BOC) ^(W) ^(X) is thein-phase or real part of TAP_(BOC) ^(W) ^(x) , and Q_TAP_(BOC) ^(W) ^(X)is the quadrature or imaginary part of TAP_(BOC) ^(W) ^(X) ; and c isthe code.
 26. The system according to claim 19 wherein the set of QBOCcorrelations is modeled by the following equation to form a secondearly-minus-late function: $\begin{matrix}{{TAP}_{QBOC}^{w_{X}} = {{AW}_{X}{\sum\limits_{{i = 0},{{di} = {F_{Chip}/F_{S}}}}^{N_{Chip}}\left\{ \begin{matrix}\begin{matrix}{c_{i}{c_{i - \tau}.}} \\{{\sin\left( {2\pi\; N_{BOC}i} \right)}{{\cos\left( {2\pi\;{N_{BOC}\left( {i - \tau} \right)}} \right)}.}}\end{matrix} \\{\mathbb{e}}^{j\;\delta\;\theta_{i}}\end{matrix} \right.}}} \\{= {\frac{A}{2}W_{X}{R(\tau)}{\sin\left( {2\pi\; N_{BOC}\tau} \right)}{\mathbb{e}}^{j\delta\varphi}}} \\{{= {\overset{\overset{{I\_ TAP}_{QBOC}^{W_{X}}}{︷}}{\frac{A}{2}W_{X}{R(\tau)}{\sin\left( {2\pi\; N_{BOC}\tau} \right)}\cos\;{\delta\varphi}} + {j\underset{\underset{{Q\_ TAP}_{QBOC}^{W_{X}}}{⎴}}{\frac{A}{2}W_{X}{R(\tau)}{\sin\left( {2\pi\; N_{BOC}\tau} \right)}\sin\;\delta\;\varphi}}}},}\end{matrix}$ where TAP_(QBOC) ^(W) ^(X) is the correlation based on theBOC input and the QBOC local signal with a selection of pseudo-noisecode (PN) TAP, where PN TAP can be early, prompt or late, i is chipphase in unit of chips, N_(Chip) is number of chips per millisecond, diis sample phase increment in unit of chips, A is the amplitude of signalat level of millisecond integration, W_(x) is the window size in chips,N_(BOC) is equal to m/n of BOC (m,n),where m is f_(m)/f_(c)and n isf_(n)/f_(c),f_(m) is a first subcarrier frequency, f_(n) is the actualchip frequency, and f_(c) is the reference chipping rate, F_(Chip) isthe chipping rate in chips/second, F_(S) is the sampling rate insamples/second, R(τ) is the correlation function for PN code, τ is thechip phase error estimated at receiver side, δθ_(i) is the carrier phaseestimation error for each sample, δφ is the average carrier phaseestimation error over N_(Chip) period, I_TAP_(QBOC) ^(W) ^(X) is thein-phase or real part of TAP_(QBOC) ^(W) ^(x) , and Q_TAP_(QBOC) ^(W)^(X) is the quadrature or imaginary part of TAP_(QBOC) ^(W) ^(X) ; and cis the code.
 27. The system according to claim 19 wherein the detectingof the primary amplitude or the secondary amplitude for the samplingperiod is based on the following equations that use linear approximationseparately for the BOC and QBOC signal components:Amp^(Approx)(Y)

max(|I_Y|,|Q_Y|)+μmin(|I_Y|,|Q_Y|), where Y + TAP_(XBOC) ^(W) ^(X) , acorrelation based on the xBOC component and the xBOC local signal with aselection of pseudo-noise code (PN) TAP, where PN TAP can be early,prompt or late, μ is a selected scale or constant scaling factor, W_(x)is the window size of a correlator, I_Y is the in-phase xBOC component,and Q_Y is the quadrature-phase xBOC component, xBOC refers to the BOCor QBOC signal component of the received composite signal.
 28. Thesystem according to claim 19 wherein the primary amplitude comprises thecombined signal power of a BOC in-phase component and a BOC quadraturephase component of the BOC component; and wherein the secondaryamplitude comprises the combined signal power of a QBOC in-phasecomponent and a QBOC quadrature-phase component of the QBOC component.29. The system according to claim 19 wherein a zero crossing point for aBOC amplitude function offsets by approximately $\frac{1}{4N_{BOC}}$against a zero crossing point for a QBOC amplitude function, where N_(BOC) is equal to m/n of BOC (m,n),where m is f_(m)/f_(c) and n isf_(n)/f_(c),f_(m) is a first subcarrier frequency, f_(n) is the actualchip frequency, and f_(c) is the reference chipping rate.
 30. The systemaccording to claim 19 further comprising: after code acquisition of thereceived signal and during a code lock mode, the tracking module isadapted to use, predominately or only, the set of BOC correlations todrive a code and carrier tracking.
 31. A method for receiving a receivedcomposite signal, the method comprising: receiving a plurality ofcomposite signals from corresponding satellites, the composite signal,from each satellite, comprising binary offset carrier (BOC) modulatedsignals that are processed to extract obtain a BOC component and toderive a quadrature BOC (QBOC) component; detecting a primary amplitudeof the BOC component of one of the received composite signals based oncorrelation data; detecting a secondary amplitude of the QBOC componentof the one of the received composite signals based on the correlationdata; determining whether the primary amplitude exceeds the secondaryamplitude for a sampling period; selecting a set of early, prompt andlate BOC correlations in accordance with a BOC correlation function forthe sampling period if the primary amplitude exceeds the secondaryamplitude for the sampling period; and selecting a set of early, prompt,and late QBOC correlations in accordance with a QBOC correlationfunction for the sampling period if the secondary amplitude exceeds theprimary amplitude for the sampling period; and processing a first subsetof the selected set of correlations to track a carrier of the one of thereceived composite signals for estimation of a range between a receiverantenna and the corresponding satellite transmitter that transmits theone of the received composite signals.
 32. The method according to claim31 wherein the first subset comprises the prompt BOC correlation or theprompt QBOC correlation.
 33. The method according to claim 32 furthercomprising: processing a second subset of the selected set ofcorrelations to form a first code error.
 34. The method according toclaim 33 wherein the second subset comprises a BOC pair of an early BOCcorrelation and a late BOC correlation, or a QBOC pair of a respectiveearly QBOC correlation and a respective late QBOC correlation, whereeach of the pairs of the correlations has a first chip spacing to drivea code tracking.